Code division multiple access (CDMA) communication system

ABSTRACT

A subscriber unit for use in a multiple access spread-spectrum communication system includes a spread spectrum radio interface, responsive to a rate function signal from a base station, and first and second despreaders. The base station assigns the rate function spread-spectrum message channels and the first despreader recovers and modifies an information signal one of the spread spectrum message channels. The information channel mode is then modified for processing by the second despreader, with the second despreader supporting a different information signal rate. The subscriber unit has a capability of communicating with a dynamically changing a transmission rate of an information signal which includes multiple spread spectrum message channels. The system includes a closed loop power control system for maintaining a minimum system transmit power level for a radio carrier station and the subscriber units, and system capacity management for maintaining a maximum number of active subscriber units for improved system performance.

[0001] This application is a continuation in part of U.S. patentapplication Ser. No. 08/669,775 filed on Jun. 27, 1996.

BACKGROUND OF THE INVENTION

[0002] The present invention generally pertains to Code DivisionMultiple Access (CDMA) communications, also known as spread-spectrumcommunications. More particularly, the present invention pertains to asystem and method for providing a high capacity, CDMA communicationssystem which provides for one or more simultaneous user bearer channelsover a given radio frequency, allowing dynamic allocation of bearerchannel rate while rejecting multipath interference.

DESCRIPTION OF THE RELEVANT ART

[0003] Providing quality telecommunication services to user-groups whichare classified as remote, such as rural telephone systems and telephonesystems in underdeveloped countries, has proved to be a challenge inrecent years. These needs have been partially satisfied by wirelessradio services, such as fixed or mobile frequency division multiplex(FDM), frequency division multiple access (FDMA), time divisionmultiplex (TDM), time division multiple access (TDMA) systems,combination frequency and time division systems (FD/TDMA), and otherland mobile radio systems. Usually, these remote services are faced withmore potential users than can be supported simultaneously by theirfrequency or spectral bandwidth capacity.

[0004] Recognizing these limitations, recent advances in wirelesscommunications have used spread spectrum modulation techniques toprovide simultaneous communication by multiple users. Spread spectrummodulation refers to modulating an information signal with a spreadingcode signal; the spreading code signal being generated by a codegenerator where the period T_(c) of the spreading code is substantiallyless than the period of the information data bit or symbol signal. Thecode may modulate the carrier frequency upon which the information hasbeen sent, called frequency-hopped spreading, or may directly modulatethe signal by multiplying the spreading code with the information datasignal, called direct-sequence spreading (DS). Spread-spectrummodulation produces a signal with bandwidth substantially greater thanthat required to transmit the information signal. Synchronous receptionand despreading of the signal at the receiver recovers the originalinformation. A synchronous demodulator in the receiver uses a referencesignal to synchronize the despreading circuits to the inputspread-spectrum modulated signal to recover the carrier and informationsignals. The reference signal can be a spreading code which is notmodulated by an information signal. Such use of a synchronousspread-spectrum modulation and demodulation for wireless communicationis described in U.S. Pat. No. 5,228,056 entitled SYNCHRONOUSSPREAD-SPECTRUM COMMUNICATIONS SYSTEM AND METHOD by Donald L. Schilling,which techniques are incorporated herein by reference.

[0005] Spread-spectrum modulation in wireless networks offers manyadvantages because multiple users may use the same frequency band withminimal interference to each user's receiver. Spread-spectrum modulationalso reduces effects from other sources of interference. In addition,synchronous spread-spectrum modulation and demodulation techniques maybe expanded by providing multiple message channels for a single user,each spread with a different spreading code, while still transmittingonly a single reference signal to the user. Such use of multiple messagechannels modulated by a family of spreading codes synchronized to apilot spreading code for wireless communication is described in U.S.Pat. No. 5,166,951 entitled HIGH CAPACITY SPREAD-SPECTRUM CHANNEL byDonald L. Schilling, which is incorporated herein by reference.

[0006] One area in which spread-spectrum techniques are used is in thefield of mobile cellular communications to provide personalcommunication services (PCS). Such systems desirably support largenumbers of users, control Doppler shift and fade, and provide high speeddigital data signals with low bit error rates. These systems employ afamily of orthogonal or quasi-orthogonal spreading codes, with a pilotspreading code sequence synchronized to the family of codes. Each useris assigned one of the spreading codes as a spreading function. Relatedproblems of such a system are: supporting a large number of users withthe orthogonal codes, handling reduced power available to remote units,and handling multipath fading effects. Solutions to such problemsinclude using phased-array antennas to generate multiple steerablebeams, using very long orthogonal or quasi-orthogonal code sequences.These sequences may be reused by cyclic shifting of the codesynchronized to a central reference, and diversity combining ofmultipath signals. Such problems associated with spread spectrumcommunications, and methods to increase capacity of a multiple access,spread-spectrum system are described in U.S. Pat. No. 4,901,307 entitledSPREAD SPECTRUM MULTIPLE ACCESS COMMUNICATION SYSTEM USING SATELLITE ORTERRESTRIAL REPEATERS by Gilhousen et al. which is incorporated hereinby reference.

[0007] The problems associated with the prior art systems focus aroundreliable reception and synchronization of the receiver despreadingcircuits to the received signal. The presence of multipath fadingintroduces a particular problem with spread spectrum receivers in that areceiver must somehow track the multipath components to maintaincode-phase lock of the receiver's despreading means with the inputsignal. Prior art receivers generally track only one or two of themultipath signals, but this method is not satisfactory because thecombined group of low power multipath signal components may actuallycontain far more power than the one or two strongest multipathcomponents. The prior art receivers track and combine the strongestcomponents to maintain a predetermined Bit Error Rate (BER) of thereceiver. Such a receiver is described, for example, in U.S. Pat. No.5,109,390 entitled DIVERSITY RECEIVER IN A CDMA CELLULAR TELEPHONESYSTEM by Gilhousen et al. A receiver that combines all multipathcomponents, however, is able to maintain the desired BER with a signalpower that is lower than that of prior art systems because more signalpower is available to the receiver. Consequently, there is a need for aspread spectrum communication system employing a receiver that trackssubstantially all of the multipath signal components, so thatsubstantially all multipath signals may be combined in the receiver, andhence the required transmit power of the signal for a given BER may bereduced.

[0008] Another problem associated with multiple access, spread-spectrumcommunication systems is the need to reduce the total transmitted powerof users in the system, since users may have limited available power. Anassociated problem requiring power control in spread-spectrum systems isrelated to the inherent characteristic of spread-spectrum systems thatone user's spread-spectrum signal is received by another user's receiveras noise with a certain power level. Consequently, users transmittingwith high levels of signal power may interfere with other users'reception. Also, if a user moves relative to another user's geographiclocation, signal fading and distortion require that the users adjusttheir transmit power level to maintain a particular signal quality. Atthe same time, the system should keep the power that the base stationreceives from all users relatively constant. Finally, because it ispossible for the spread-spectrum system to have more remote users thancan be supported simultaneously, the power control system should alsoemploy a capacity management method which rejects additional users whenthe maximum system power level is reached.

[0009] Prior spread-spectrum systems have employed a base station thatmeasures a received signal and sends an adaptive power control (APC)signal to the remote users. Remote users include a transmitter with anautomatic gain control (AGC) circuit which responds to the APC signal.In such systems the base station monitors the overall system power orthe power received from each user, and sets the APC signal accordingly.Such a spread-spectrum power control system and method is described inU.S. Pat. No. 5,299,226 entitled ADAPTIVE POWER CONTROL FOR A SPREADSPECTRUM COMMUNICATION SYSTEM AND METHOD, and U.S. Pat. No. 5,093,840entitled ADAPTIVE POWER CONTROL FOR A SPREAD SPECTRUM TRANSMITTER, bothby Donald L. Schilling and incorporated herein by reference. This openloop system performance may be improved by including a measurement ofthe signal power received by the remote user from the base station, andtransmitting an APC signal back to the base station to effectuate aclosed loop power control method. Such closed loop power control isdescribed, for example, in U.S. Pat. No. 5,107,225 entitled HIGH DYNAMICRANGE CLOSED LOOP AUTOMATIC GAIN CONTROL CIRCUIT to Charles E. Wheatley,III et al. and incorporated herein by reference.

[0010] These power control systems, however, exhibit severaldisadvantages. First, the base station must perform complex powercontrol algorithms, increasing the amount of processing in the basestation. Second, the system actually experiences several types of powervariation: variation in the noise power caused by the variation in thenumber of users and variations in the received signal power of aparticular bearer channel. These variations occur with differentfrequency, so simple power control algorithms can be optimized tocompensate for only one of the two types of variation. Finally, thesepower algorithms tend to drive the overall system power to a relativelyhigh level. Consequently, there is a need for a spread-spectrum powercontrol method that rapidly responds to changes in bearer channel powerlevels, while simultaneously making adjustments to all users' transmitpower in response to changes in the number of users. Also, there is aneed for an improved spread-spectrum communication system employing aclosed loop power control system which minimizes the system's overallpower requirements while maintaining a sufficient BER at the individualremote receivers. In addition, such a system should control the initialtransmit power level of a remote user and manage total system capacity.

[0011] Spread-spectrum communication systems desirably should supportlarge numbers of users, each of which has at least one communicationchannel. In addition, such a system should provide multiple genericinformation channels to broadcast information to all users and to enableusers to gain access to the system. Using prior art spread-spectrumsystems this could only be accomplished by generating large numbers ofspreading code sequences.

[0012] Further, spread-spectrum systems should use sequences that areorthogonal or nearly orthogonal to reduce the probability that areceiver locks to the wrong spreading code sequence or phase. The use ofsuch orthogonal codes and the benefits arising therefrom are outlined inU.S. Pat. No. 5,103,459 entitled SYSTEM AND METHOD FOR GENERATING SIGNALWAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM, by Gilhousen et al. andU.S. Pat. No. 5,193,094 entitled METHOD AND APPARATUS FOR GENERATINGSUPER-ORTHOGONAL CONVOLUTIONAL CODES AND THE DECODING THEREOF, by AndrewJ. Viterbi, both of which are incorporated herein by reference. However,generating such large families of code sequences with such properties isdifficult. Also, generating large code families requires generatingsequences which have a long period before repetition. Consequently, thetime a receiver takes to achieve synchronization with such a longsequence is increased. Prior art spreading code generators often combineshorter sequences to make longer sequences, but such sequences may nolonger be sufficiently orthogonal. Therefore, there is a need for animproved method for reliably generating large families of code sequencesthat exhibit nearly orthogonal characteristics and have a long periodbefore repetition, but also include the benefit of a short code sequencethat reduces the time to acquire and lock the receiver to the correctcode phase. In addition, the code generation method should allowgeneration of codes with any period, since the spreading code period isoften determined by parameters used such as data rate or frame size.

[0013] Another desirable characteristic of spreading code sequences isthat the transition of the user data value occur at a transition of thecode sequence values. Since data typically has a period which isdivisible by 2^(N), such a characteristic usually requires thecode-sequence to be an even length of 2^(N). However, code generators,as is well known in the art, generally use linear feedback shiftregisters which generate codes of length 2^(N)−1. Some generatorsinclude a method to augment the generated code sequence by inserting anadditional code value, as described, for example, in U.S. Pat. No.5,228,054 entitled POWER-OF-TWO LENGTH PSEUDONOISE SEQUENCE GENERATORWITH FAST OFFSET ADJUSTMENT by Timothy Rueth et al and incorporatedherein by reference. Consequently, the spread-spectrum communicationsystem should also generate spreading code sequences of even length.

[0014] Finally, the spread-spectrum communication system should be ableto handle many different types of data, such as FAX, voiceband data, andISDN, in addition to traditional voice traffic. To increase the numberof users supported, many systems employ encoding techniques such asADPCM to achieve “compression” of the digital telephone signal. FAX,ISDN and other data, however, require the channel to be a clear channel.Consequently, there is a need for a spread spectrum communication systemthat supports compression techniques that also dynamically modify thespread spectrum bearer channel between an encoded channel and a clearchannel in response to the type of information contained in the user'ssignal.

SUMMARY OF THE INVENTION

[0015] The present invention is embodied in a multiple access,spread-spectrum communication system which processes a plurality ofinformation signals received simultaneously over telecommunication linesfor simultaneous transmission over a radio frequency (RF) channel as acode-division-multiplexed (CDM) signal. The system includes a radiocarrier station (RCS) which receives a call request signal thatcorresponds to a telecommunication line information signal, and a useridentification signal that identifies a user to which the call requestand information signal are addressed. The receiving apparatus is coupledto a plurality of code division multiple access (CDMA) modems, one ofwhich provides a global pilot code signal and a plurality of messagecode signals, and each of the CDMA modems combines one of the pluralityof information signals with its respective message code signal toprovide a spread-spectrum processed signal. The plurality of messagecode signals of the plurality of CDMA modems are synchronized to theglobal pilot code signal. The system also includes assignment apparatusthat is responsive to a channel assignment signal for coupling therespective information signals received on the telecommunication linesto indicated ones of the plurality of modems; The assignment apparatusis coupled to a time-slot exchange means. The system further includes asystem channel controller coupled to a remote call-processor and to thetime-slot exchange means. The system channel controller is responsive tothe user identification signal, to provide the channel assignmentsignal. In the system, an RF transmitter is connected to all of themodems to combine the plurality of spread-spectrum processed messagesignals with the global pilot code signal to generate a CDM signal. TheRF transmitter also modulates a carrier signal with the CDM signal andtransmits the modulated carrier signal through an RF communicationchannel.

[0016] The transmitted CDM signal is received from the RF communicationchannel by a subscriber unit (SU) which processes and reconstructs thetransmitted information signal assigned to the subscriber. The SUincludes a receiving means for receiving and demodulating the CDM signalfrom the carrier. In addition, the SU comprises a subscriber unitcontroller and a CDMA modem which includes a processing means foracquiring the global pilot code and despreading the spread-spectrumprocessed signal to reconstruct the transmitted information signal.

[0017] The RCS and the SUs each contain CDMA modems for transmission andreception of telecommunication signals including information signals andconnection control signals. The CDMA modem comprises a modem transmitterhaving: a code generator for providing an associated pilot code signaland for generating a plurality of message code signals; a spreadingmeans for combining each of the information signals, with a respectiveone of the message code signals to generate spread-spectrum processedmessage signals; and a global pilot code generator which provides aglobal pilot code signal to which the message code signals aresynchronized.

[0018] The CDMA modem also comprises a modem receiver having associatedpilot code acquisition and tracking logic. The associated pilot codeacquisition logic includes an associated pilot code generator; a groupof associated pilot code correlators for correlating code-phase delayedversions of the associated pilot signal with a receive CDM signal forproducing a despread associated pilot signal. The code phase of theassociated pilot signal is changed responsive to an acquisition signalvalue until a detector indicates the presence of the despread associatedpilot code signal by changing the acquisition signal value. Theassociated pilot code signal is synchronized to the global pilot signal.The associated pilot code tracking logic adjusts the associated pilotcode signal in phase responsive to the acquisition signal so that thesignal power level of the despread associated pilot code signal ismaximized. Finally, the CDMA modem receiver includes a group of messagesignal acquisition circuits. Each message signal acquisition circuitincludes a plurality of receive message signal correlators forcorrelating one of the local receive message code signals with the CDMsignal to produce a respective despread receive message signal.

[0019] To generate large families of nearly mutually orthogonal codesused by the CDMA modems, the present invention includes a code sequencegenerator. The code sequences are assigned to a respective logicalchannel of the spread-spectrum communication system, which includesIn-phase (I) and Quadrature (Q) transmission over RF communicationchannels. One set of sequences is used as pilot sequences which are codesequences transmitted without modulation by a data signal. The codesequence generator circuit includes a long code sequence generatorincluding a linear feedback shift register, a memory which provides ashort, even code sequence, and a plurality of cyclic shift, feedforwardsections which provide other members of the code family which exhibitminimal correlation with the code sequence applied to the feedforwardcircuit. The code sequence generator further includes a group of codesequence combiners for combining each phase shifted version of the longcode sequence with the short, even code sequence to produce a group, orfamily, of nearly mutually orthogonal codes.

[0020] Further, the present invention includes several methods forefficient utilization of the spread-spectrum channels. First, the systemincludes a bearer channel modification system which comprises a group ofmessage channels between a first transceiver and second transceiver.Each of the group of message channels supports a different informationsignal transmission rate. The first transceiver monitors a receivedinformation signal to determine the type of information signal that isreceived, and produces a coding signal relating to the coding signal. Ifa certain type of information signal is present, the first transceiverswitches transmission from a first message channel to a second messagechannel to support the different transmission rate. The coding signal istransmitted by the first transceiver to the second transceiver, and thesecond transceiver switches to the second message channel to receive theinformation signal at a different transmission rate.

[0021] Another method to increase efficient utilization of the bearermessage channels is the method of idle-code suppression used by thepresent invention. The spread-spectrum transceiver receives a digitaldata information signal including a predetermined flag patterncorresponding to an idle period. The method includes the steps of: 1)delaying and monitoring the digital data signal; 2) detecting thepredetermined flag pattern; 3) suspending transmission of the digitaldata signal when the flag pattern is detected; and 4) transmitting thedata signal as a spread-spectrum signal when the flag pattern is notdetected.

[0022] The present invention includes a system and method for closedloop automatic power control (APC) for the RCS and SUs of thespread-spectrum communication system. The SUs transmit spread-spectrumsignals, the RCS acquires the spread-spectrum signals, and the RCSdetects the received power level of the spread-spectrum signals plus anyinterfering signal including noise. The APC system includes the RCS anda plurality of SUs, wherein the RCS transmits a plurality of forwardchannel information signals to the SUs as a plurality of forward channelspread-spectrum signals having a respective forward transmit powerlevel, and each SU transmits to the base station at least one reversespread-spectrum signal having a respective reverse transmit power leveland at least one reverse channel spread-spectrum signal which includes areverse channel information signal.

[0023] The APC includes an automatic forward power control (AFPC)system, and an automatic reverse power control (ARPC) system. The AFPCsystem operates by measuring, at the SU, a forward signal-to-noise ratioof the respective forward channel information signal, generating arespective forward channel error signal corresponding to a forward errorbetween the respective forward signal-to-noise ratio and apre-determined signal-to-noise value, and transmitting the respectiveforward channel error signal as part of a respective reverse channelinformation signal from the SU to the RCS. The RCS includes a pluralnumber of AFPC receivers for receiving the reverse channel informationsignals and extracting the forward channel error signals from therespective reverse channel information signals. The RCU also adjusts therespective forward transmit power level of each one of the respectiveforward spread-spectrum signals responsive to the respective forwarderror signal.

[0024] The ARPC system operates by measuring, in the RCS, a reversesignal-to-noise ratio of each of the respective reverse channelinformation signals, generating a respective reverse channel errorsignal representing an error between the respective reverse channelsignal-to-noise ratio and a respective pre-determined signal-to-noisevalue, and transmitting the respective reverse channel error signal as apart of a respective forward channel information signal to the SU. EachSU includes an ARPC receiver for receiving the forward channelinformation signal and extracting the respective reverse error signalfrom the forward channel information signal. The SU adjusts the reversetransmit power level of the respective reverse spread-spectrum signalresponsive to the respective reverse error signal.

BRIEF DESCRIPTION OF THE DRAWINGS

[0025]FIG. 1 is a block diagram of a code division multiple accesscommunication system according to the present invention.

[0026]FIG. 2a is a block diagram of a 36 stage linear shift registersuitable for use with long spreading code of the code generator of thepresent invention.

[0027]FIG. 2b is a block diagram of circuitry which illustrates thefeed-forward operation of the code generator.

[0028]FIG. 2c is a block diagram of an exemplary code generator of thepresent invention including circuitry for generating spreading codesequences from the long spreading codes and the short spreading codes.

[0029]FIG. 2d is an alternate embodiment of the code generator circuitincluding delay elements to compensate for electrical circuit delays.

[0030]FIG. 3a is a graph of the constellation points of the pilotspreading code QPSK signal.

[0031]FIG. 3b is a graph of the constellation points of the messagechannel QPSK signal.

[0032]FIG. 3c is a block diagram of exemplary circuitry which implementsthe method of tracking the received spreading code phase of the presentinvention.

[0033]FIG. 3d is a block diagram of an alternative exemplary circuitrywhich implements the method of tracking the received spreading codephase of the present invention.

[0034]FIG. 4 is a block diagram of the tracking circuit that tracks themedian of the received multipath signal components.

[0035]FIG. 5a is a block diagram of the tracking circuit that tracks thecentroid of the received multipath signal components.

[0036]FIG. 5b is a block diagram of the Adaptive Vector Correlator.

[0037]FIG. 6 is a block diagram of exemplary circuitry which implementsthe acquisition decision method of the correct spreading code phase ofthe received pilot code of the present invention.

[0038]FIG. 7 is a block diagram of an exemplary pilot rake filter whichincludes the tracking circuit and digital phase locked loop fordespreading the pilot spreading code, and generator of the weightingfactors of the present invention.

[0039]FIG. 8a is a block diagram of an exemplary adaptive vectorcorrelator and matched filter for despreading and combining themultipath components of the present invention.

[0040]FIG. 8b is a block diagram of an alternative implementation of theadaptive vector correlator and adaptive matched filter for despreadingand combining the multipath components of the present invention.

[0041]FIG. 8c is a block diagram of an alternative embodiment of theadaptive vector correlator and adaptive matched filter for despreadingand combining the multipath components of the present invention.

[0042]FIG. 8d is a block diagram of the Adaptive Matched Filter of oneembodiment of the present invention.

[0043]FIG. 9 is a block diagram of the elements of an exemplary radiocarrier station (RCS) of the present invention.

[0044]FIG. 10 is a block diagram of the elements of an exemplarymultiplexer suitable for use in the RCS shown in FIG. 9.

[0045]FIG. 11 is a block diagram of the elements of an exemplarywireless access controller (WAC) of the RCS shown in FIG. 9.

[0046]FIG. 12 is a block diagram of the elements of an exemplary modeminterface unit (MIU) of the RCS shown in FIG. 9.

[0047]FIG. 13 is a high level block diagram showing the transmit,receive, control, and code generation circuitry of the CDMA modem.

[0048]FIG. 14 is a block diagram of the transmit section of the CDMAmodem.

[0049]FIG. 15 is a block diagram of an exemplary modem input signalreceiver.

[0050]FIG. 16 is a block diagram of an exemplary convolutional encoderas used in the present invention.

[0051]FIG. 17 is a block diagram of the receive section of the CDMAmodem.

[0052]FIG. 18 is a block diagram of an exemplary adaptive matched filteras used in the CDMA modem receive section.

[0053]FIG. 19 is a block diagram of an exemplary pilot rake as used inthe CDMA modem receive section.

[0054]FIG. 20 is a block diagram of an exemplary auxiliary pilot rake asused in the CDMA modem receive section.

[0055]FIG. 21 is a block diagram of an exemplary video distributioncircuit (VDC) of the RCS shown in FIG. 9.

[0056]FIG. 22 is a block diagram of an exemplary RF transmitter/receiverand exemplary power amplifiers of the RCS shown in FIG. 9.

[0057]FIG. 23 is a block diagram of an exemplary subscriber unit (SU) ofthe present invention.

[0058]FIG. 24 is a flow-chart diagram of an exemplary call establishmentalgorithm for an incoming call request used by the present invention forestablishing a bearer channel between an RCS and an SU.

[0059]FIG. 25 is a flow-chart diagram of an exemplary call establishmentalgorithm for an outgoing call request used by the present invention forestablishing a bearer channel between an RCS and an SU.

[0060]FIG. 26 is a flow-chart diagram of an exemplary maintenance powercontrol algorithm of the present invention.

[0061]FIG. 27 is a flow-chart diagram of an exemplary automatic forwardpower control algorithm of the present invention.

[0062]FIG. 28 is a flow-chart diagram of an exemplary automatic reversepower control algorithm of the present invention.

[0063]FIG. 29 is a block diagram of an exemplary closed loop powercontrol system of the present invention when the bearer channel isestablished.

[0064]FIG. 30 is a block diagram of an exemplary closed loop powercontrol system of the present invention during the process ofestablishing the bearer channel.

[0065]FIG. 31 is a diagram of the RCS and SU configured for testpurposes.

GLOSSARY OF ACRONYMS

[0066] Acronym Definition AC Assigned Channels A/D Analog-to-DigitalADPCM Adaptive Differential Pulse Code Modulation AFPC Automatic ForwardPower Control AGC Automatic Gain Control AMF Adaptive Matched Filter APCAutomatic Power Control ARPC Automatic Reverse Power Control ASPTAssigned Pilot AVC Adaptive Vector Correlator AXCH Access Channel B-CDMABroadband Code Division Multiple Access BCM Bearer Channel ModificationBER Bit Error Rate BS Base Station CC Call Control CDM Code DivisionMultiplex CDMA Code Division Multiple Access CLK Clock Signal GeneratorCO Central Office CTCH Control Channel CUCH Check-Up Channel dB DecibelsDCC Data Combiner Circuitry DI Distribution Interface DLL Delay LockedLoop DM Delta Modulator DS Direct Sequence EPIC Extended PCM InterfaceController FBCH Fast Broadcast Channel FDM Frequency Division MultiplexFD/TDMA Frequency & Time Division Systems FDMA Frequency DivisionMultiple Access FEC Forward Error Correction FSK Frequency Shift KeyingFSU Fixed Subscriber Unit GC Global Channel GLPT Global Pilot GPC GlobalPilot Code GPSK Gaussian Phase Shift Keying GPS Global PositioningSystem HPPC High Power Passive Components HSB High Speed Bus I In-PhaseIC Interface Controller ISDN Integrated Services Digital Network ISSTInitial System Signal Threshold LAXPT Long Access Pilot LAPD Link AccessProtocol LCT Local Craft Terminal LE Local Exchange LFSR Linear FeedbackShift Register LI Line Interface LMS Least Mean Square LOL Loss of CodeLock LPF Low Pass Filter LSR Linear Shift Register MISR Modem InputSignal Receiver MIU Modem Interface Unit MM Mobility Management MOIModem Output Interface MPC Maintenance Power Control MPSK M-ary PhaseShift Keying MSK Minimum Shift Keying MSU Mobile Subscriber Unit NENetwork Element OMS Operation and Maintenance System OS OperationsSystem OQPSK Offset Quadrature Phase Shift Keying OW Order Wire PARKPortable Access Rights Key PBX Private Branch Exchange PCM Pulse CodedModulation PCS Personal Communication Services PG Pilot Generator PLLPhase Locked Loop PLT Pilot PN Pseudonoise POTS Plain Old TelephoneService PSTN Public Switched Telephone Network Q Quadrature QPSKQuadrature Phase Shift Keying RAM Random Access Memory RCS Radio CarrierStation RDI Receiver Data Input Circuit RDU Radio Distribution Unit RFRadio Frequency RLL Radio Local Loop SAXPT Short Access Channel PilotsSBCH Slow Broadcast Channel SHF Super High Frequency SIR Signal Power toInterface Noise Power Ratio SLIC Subscriber Line Interface Circuit SNRSignal-to-Noise Ratio SPC Service PC SPRT Sequential Probability RatioTest STCH Status Channel SU Subscriber Unit TDM Time DivisionMultiplexing TMN Telecommunication Management Network TRCH TrafficChannels TSI Time-Slot Interchanger TX Transmit TXIDAT I-Modem TransmitData Signal TXQDAT Q-Modem Transmit Data Signal UHF Ultra High FrequencyVCO Voltage Controlled Oscillator VDC Video Distribution Circuit VGAVariable Gain Amplifier VHF Very High Frequency WAC Wireless AccessController

DESCRIPTION OF THE EXEMPLARY EMBODIMENT

[0067] General System Description

[0068] The system of the present invention provides local-loop telephoneservice using radio links between one or more base stations and multipleremote subscriber units. In the exemplary embodiment, a radio link isdescribed for a base station communicating with a fixed subscriber unit(FSU), but the system is equally applicable to systems includingmultiple base stations with radio links to both FSUs and MobileSubscriber Units (MSUs). Consequently, the remote subscriber units arereferred to herein as Subscriber Units (SUs).

[0069] Referring to FIG. 1, Base Station (BS) 101 provides callconnection to a local exchange (LE) 103 or any other telephone networkswitching interface, such as a private branch exchange (PBX) andincludes a Radio Carrier Station (RCS) 104. One or more RCSs 104, 105,110 connect to a Radio Distribution Unit (RDU) 102 through links 131,132, 137, 138, 139, and RDU 102 interfaces with LE 103 by transmittingand receiving call set-up, control, and information signals throughtelco links 141, 142, 150. SUs 116, 119 communicate with the RCS 104through radio links 161, 162, 163, 164, 165. Alternatively, anotherembodiment of the invention includes several SUs and a “master” SU withfunctionality similar to the RCS. Such an embodiment may or may not haveconnection to a local telephone network.

[0070] The radio links 161 to 165 operate within the frequency bands ofthe DCS1800 standard (1.71-1.785 Ghz and 1.805-1.880 GHz); the US-PCSstandard (1.85-1.99 Ghz); and the CEPT standard (2.0-2.7 GHz). Althoughthese bands are used in described embodiment, the invention is equallyapplicable to the entire UHF to SHF bands, including bands from 2.7 GHzto 5 GHz. The transmit and receive bandwidths are multiples of 3.5 MHzstarting at 7 MHz, and multiples of 5 MHz starting at 10 MHz,respectively. The described system includes bandwidths of 7, 10, 10.5,14 and 15 MHz. In the exemplary embodiment of the invention, the minimumguard band between the Uplink and Downlink is 20 MHz, and is desirablyat least three times the signal bandwidth. The duplex separation isbetween 50 to 175 MHz, with the described invention using 50, 75, 80,95, and 175 MHz. Other frequencies may also be used.

[0071] Although the described embodiment uses different spread-spectrumbandwidths centered around a carrier for the transmit and receivespread-spectrum channels, the present method is readily extended tosystems using multiple spread-spectrum bandwidths for the transmitchannels and multiple spread-spectrum bandwidths for the receivechannels. Alternatively, because spread-spectrum communication systemshave the inherent feature that one user's transmission appears as noiseto another user's despreading receiver, an embodiment may employ thesame spread-spectrum channel for both the transmit and receive pathchannels. In other words, Uplink and Downlink transmissions can occupythe same frequency band. Furthermore, the present method may be readilyextended to multiple CDMA frequency bands, each conveying a respectivelydifferent set of messages, uplink, downlink or uplink and downlink.

[0072] The spread binary symbol information is transmitted over theradio links 161 to 165 using Quadrature Phase Shift Keying (QPSK)modulation with Nyquist Pulse Shaping in the present embodiment,although other modulation techniques may be used, including, but notlimited to, Offset QPSK (OQPSK) and Minimum Shift Keying (MSK). GaussianPhase Shift Keying (GPSK) and M-ary Phase Shift Keying (MPSK) The radiolinks 161 to 165 incorporate Broadband Code Division Multiple Access(B-CDMA™) as the mode of transmission in both the Uplink and Downlinkdirections. CDMA (also known as Spread Spectrum) communicationtechniques used in multiple access systems are well-known, and aredescribed in U.S. Pat. No. 5,228,056 entitled SYNCHRONOUSSPREAD-SPECTRUM COMMUNICATION SYSTEM AND METHOD by Donald T Schilling.The system described utilizes the Direct Sequence (DS) spreadingtechnique. The CDMA modulator performs the spread-spectrum spreadingcode sequence generation, which can be a pseudonoise (PN) sequence; andcomplex DS modulation of the QPSK signals with spreading code sequencesfor the In-phase (I) and Quadrature (Q) channels. Pilot signals aregenerated and transmitted with the modulated signals, and pilot signalsof the present embodiment are spreading codes not modulated by data. Thepilot signals are used for synchronization, carrier phase recovery, andfor estimating the impulse response of the radio channel. Each SUincludes a single pilot generator and at least one CDMA modulator anddemodulator, together known as a CDMA modem. Each RCS 104, 105, 110 hasa single pilot generator plus sufficient CDMA modulators anddemodulators for all of the logical channels in use by all SUs.

[0073] The CDMA demodulator despreads the signal with appropriateprocessing to combat or exploit multipath propagation effects.Parameters concerning the received power level are used to generate theAutomatic Power Control (APC) information which, in turn, is transmittedto the other end of the communication link. The APC information is usedto control transmit power of the automatic forward power control (AFPC)and automatic reverse power control (ARPC) links. In addition, each RCS104, 105 and 110 can perform Maintenance Power Control (MPC), in amanner similar to APC, to adjust the initial transmit power of each SU111, 112, 115, 117 and 118. Demodulation is coherent where the pilotsignal provides the phase reference.

[0074] The described radio links support multiple traffic channels withdata rates of 8, 16, 32, 64, 128, and 144 kb/s. The physical channel towhich a traffic channel is connected operates with a 64 k symbol/secrate. Other data rates may be supported, and Forward Error Correction(FEC) coding can be employed. For the described embodiment, FEC withcoding rate of ½ and constraint length 7 is used. Other rates andconstraint lengths can be used consistent with the code generationtechniques employed.

[0075] Diversity combining at the radio antennas of RCS 104, 105 and 110is not necessary because CDMA has inherent frequency diversity due tothe spread bandwidth. Receivers include Adaptive Matched Filters (AMFs)(not shown in FIG. 1) which combine the multipath signals. In thepresent embodiment, the exemplary AMFs perform Maximal Ratio Combining.

[0076] Referring to FIG. 1, RCS 104 interfaces to RDU 102 through links131, 132, 137 with, for example, 1.544 Mb/s DS1, 2.048 Mb/s E1; or HDSLFormats to receive and send digital data signals. While these aretypical telephone company standardized interfaces, the present inventionis not limited to these digital data formats only. The exemplary RCSline interface (not shown in FIG. 1) translates the line coding (such asHDB3, B8ZS, AMI) and extracts or produces framing information, performsAlarms and Facility signaling functions, as well as channel specificloop-back and parity check functions. The interfaces for thisdescription provide 64 kb/s PCM encoded or 32 kb/s ADPCM encodedtelephone traffic channels or ISDN channels to the RCS for processing.Other ADPCM encoding techniques can be used consistent with the sequencegeneration techniques.

[0077] The system of the present invention also supports bearer ratemodification between the RCS 104 and each SU 111, 112, 115, 117 and 118communicating with RCS 104 in which a CDMA message channel supporting 64kb/s may be assigned to voiceband data or FAX when rates above 4.8 kb/sare present. Such 64 kb/s bearer channel is considered an unencodedchannel. For ISDN, bearer rate modification may be done dynamically,based upon the D channel messages.

[0078] In FIG. 1, each SU 111, 112, 115, 117 and 118 either includes orinterfaces with a telephone unit 170, or interfaces with a local switch(PBX) 171. The input from the telephone unit may include voice,voiceband data and signaling. The SU translates the analog signals intodigital sequences, and may also include a Data terminal 172 or an ISDNinterface 173. The SU can differentiate voice input, voiceband data orFAX and digital data. The SU encodes voice data with techniques such asADPCM at 32 kb/s or lower rates, and detects voiceband data or FAX withrates above 4.8 kb/s to modify the traffic channel (bearer ratemodification) for unencoded transmission. Also, A-law, u-law, or nocompanding of the signal may be performed before transmission. Fordigital data, data compression techniques, such as idle flag removal,may also be used to conserve capacity and minimize interference.

[0079] The transmit power levels of the radio interface between RCS 104and SUs 111, 112, 115, 117 and 118 are controlled using two differentclosed loop power control methods. The Automatic Forward Power Control(AFPC) method determines the Downlink transmit power level, and theAutomatic Reverse Power Control (ARPC) method determines the Uplinktransmit power level. The logical control channel by which SU 111 andRCS 104, for example; transfer power control information operates atleast a 16 kHz update rate. Other embodiments may use a faster or slowerupdate rate, for example at 64 kHz or 8 kHz. These algorithms ensurethat the transmit power of a user maintains an acceptable Bit-Error Rate(BER), maintains the system power at a minimum to conserve power, andmaintains the power level of all SUs 111, 112, 115, 117 and 118 receivedby RCS 104 at a nearly equal level.

[0080] In addition, the system uses an optional maintenance powercontrol method during the inactive mode of a SU. When SU 111 is inactiveor powered-down to conserve power, the unit occasionally activates toadjust its initial transmit power level setting in response to amaintenance power control signal from RCS 104. The maintenance powersignal is determined by the RCS 104 by measuring the received powerlevel of SU 111 and present system power level and, from this,calculates the necessary initial transmit power. The method shortens thechannel acquisition time of SU 111 to begin a communication. The methodalso prevents the transmit power level of SU 111 from becoming too highand interfering with other channels during the initial transmissionbefore the closed loop power control reduces the transmit power.

[0081] RCS 104 obtains synchronization of its clock from an interfaceline such as, but not limited to, E1, T1, or HDSL interfaces. RCS 104can also generate its own internal clock signal from an oscillator whichmay be regulated by a Global Positioning System (GPS) receiver. RCS 104generates a Global Pilot Code, a channel with a spreading code but nodata modulation, which can be acquired by remote SUs 111 through 118.All transmission channels of the RCS are synchronized to the Pilotchannel, and spreading code phases of code generators (not shown) usedfor Logical communication channels within RCS 104 are also synchronizedto the Pilot channel's spreading code phase. Similarly, SUs 111 through118 which receive the Global Pilot Code of RCS 104 synchronize thespreading and de-spreading code phases of the code generators (notshown) of the SUs to the Global Pilot Code.

[0082] RCS 104, SU 111, and RDU 102 may incorporate system redundancy ofsystem elements and automatic switching between internal functionalsystem elements upon a failure event to prevent loss or drop-out of aradio link, power supply, traffic channel, or group of traffic channels.

[0083] Logical Communication Channels

[0084] A ‘channel’ of the prior art is usually regarded as acommunications path which is part of an interface and which can bedistinguished from other paths of that interface without regard to itscontent. However, in the case of CDMA, separate communications paths aredistinguished only by their content. The term ‘logical channel’ is usedto distinguish the separate data streams, which are logically equivalentto channels in the conventional sense. All logical channels andsub-channels of the present invention are mapped to a common 64kilo-symbols per second (ksym/s) QPSK stream. Some channels aresynchronized to associated pilot codes which are generated from, andperform a similar function to the system Global Pilot Code (GPC). Thesystem pilot signals are not, however, considered logical channels.

[0085] Several logical communication channels are used over the RFcommunication link between the RCS and SU. Each logical communicationchannel either has a fixed, predetermined spreading code or adynamically assigned spreading code. For both predetermined and assignedcodes, the code phase is synchronized with the Pilot Code. Logicalcommunication channels are divided into two groups: the Global Channel(GC) group includes channels which are either transmitted from the basestation RCS to all remote SUs or from any SU to the RCS of the basestation regardless of the SU's identity. The channels in the GC groupmay contain information of a given type for all users including thosechannels used by SUs to gain system access. Channels in the AssignedChannels (AC) group are those channels dedicated to communicationbetween the RCS and a particular SU.

[0086] The Global Channels (GC) group provides for 1) Broadcast Controllogical channels, which provide point to multipoint services forbroadcasting messages to all SUs and paging messages to SUs; and 2)Access Control logical channels which provide point-to-point services onglobal channels for SUs to access the system and obtain assignedchannels.

[0087] The RCS of the present invention has multiple Access Controllogical channels, and one Broadcast Control group. An SU of the presentinvention has at least one Access Control channel and at least oneBroadcast Control logical channel.

[0088] The Global logical channels controlled by the RCS are the FastBroadcast Channel (FBCH) which broadcasts fast changing informationconcerning which services and which access channels are currentlyavailable, and the Slow Broadcast Channel (SBCH) which broadcasts slowchanging system information and paging messages. The Access Channel(AXCH) is used by the SUs to access an RCS and gain access to assignedchannels. Each AXCH is paired with a Control Channel (CTCH). The CTCH isused by the RCS to acknowledge and reply to access attempts by SUs. TheLong Access Pilot (LAXPT) is transmitted synchronously with AXCH toprovide the RCS with a time and phase reference.

[0089] An Assigned Channel (AC) group contains the logical channels thatcontrol a single telecommunication connection between the RCS and an SU.The functions developed when an AC group is formed include a pair ofpower control logical message channels for each of the Uplink andDownlink connections, and depending on the type of connection, one ormore pairs of traffic channels. The Bearer Control function performs therequired forward error control, bearer rate modification, and encryptionfunctions.

[0090] Each SU 111, 112, 115, 117 and 118 has at least one AC groupformed when a telecommunication connection exists, and each RCS 104, 105and 110 has multiple AC groups formed, one for each connection inprogress. An AC group of logical channels is created for a connectionupon successful establishment of the connection. The AC group includesencryption, FEC coding, and multiplexing on transmission, and FECdecoding, decryption and demultiplexing on reception.

[0091] Each AC group provides a set of connection orientedpoint-to-point services and operates in both directions between aspecific RCS, for example, RCS 104 and a specific SU, for example, SU111. An AC group formed for a connection can control more than onebearer over the RF communication channel associated with a singleconnection. Multiple bearers are used to carry distributed data such as,but not limited to, ISDN. An AC group can provide for the duplication oftraffic channels to facilitate switch over to 64 kb/s PCM for high speedfacsimile and modem services for the bearer rate modification function.

[0092] The assigned logical channels formed upon a successful callconnection and included in the AC group are a dedicated signalingchannel [order wire (OW)], an APC channel, and one or more Trafficchannels (TRCH) which are bearers of 8, 16, 32, pr 64 kb/s depending onthe service supported. For voice traffic, moderate rate coded speech,ADPCM, or PCM can be supported on the Traffic channels. For ISDN servicetypes, two 64 kb/s TRCHs form the B channels and a 16 kb/s TRCH formsthe D channel. Alternatively, the APC subchannel may either beseparately modulated on its own CDMA channel, or may be time divisionmultiplexed with a traffic channel or OW channel.

[0093] Each SU 111, 112, 115, 117 and 118 of the present inventionsupports up to three simultaneous traffic channels. The mapping of thethree logical channels for TRCHs to the user data is shown below inTable 1: TABLE 1 Mapping of service types to the three available TRCHchannels Service TRCH(0) TRCH(1) TRCH(2) 16 kb/s POTS TRCH/16 not usednot used 32 + 64 kb/s POTS (during BCM) TRCH/32 TRCH/64 not used 32 kb/sPOTS TRCH/32 not used not used 64 kb/s POTS not used TRCH/64 not usedISDN D not used not used TRCH/16 ISDN B + D TRCH/64 not used TRCH/16ISDN 2B + D TRCH/64 TRCH/64 TRCH/16 Digital LL @ 64 kb/s TRCH/64 notused not used Digital LL @ 2 × 64 kb/s TRCH/64 TRCH/64 not used AnalogLL @ 64 kb/s TRCH/64 not used not used

[0094] The APC data rate is sent at 64 kb/s. The APC logical channel isnot FEC coded to avoid delay and is transmitted at a relatively lowpower level to minimize capacity used for APC. Alternatively, the APCand OW may be separately modulated using complex spreading codesequences, or they may be time division multilplexed.

[0095] The OW logical channel is FEC coded with a rate ½ convolutionalcode. This logical channel is transmitted in bursts when signaling datais present to reduce interference. After an idle period, the OW signalbegins with at least 35 symbols prior to the start of the data frame.For silent maintenance call data, the OW is transmitted continuouslybetween frames of data. Table 2 summarizes the logical channels used inthe exemplary embodiment: TABLE 2 Logical Channels and sub-channels ofthe B-CDMA Air Interface Direction (forward Channel Brief or Bit Maxname Abbr. Description reverse) rate BER Power level Pilot GlobalChannels Fast FBCH Broadcasts F 16 kb/s 1e−4 Fixed GLPT Broadcastfast-changing Channel system information Slow SBCH Broadcasts F 16 kb/s1e−7 Fixed GLPT Broadcast paging Channel messages to FSUs andslow-changing system information Access AXCH(i) For initial R 32 kb/s1e−7 Controlled LAXPT(i) Channels access attempts by APC by FSUs ControlCTCH(i) For granting F 32 kb/s 1e−7 Fixed GLPT Channels access AssignedChannels 16 kb/s TRCH/ General POTS F/R 16 kb/s 1e−4 Controlled F-GLPTPOTS 16 use by APC R-ASPT 32 kb/s TRCH/ General POTS F/R 32 kb/s 1e−4Controlled F-GLPT POTS 32 use by APC R-ASPT 64 kb/s TRCH/ POTS use forF/R 64 kb/s 1e−4 Controlled F-GLPT POTS 64 in-band by APC R-ASPTmodems/fax D channel TRCH/ ISDN D F/R 16 kb/s 1e−7 Controlled F-GLPT 16channel by APC R-ASPT Order OW assigned F/R 32 kb/s 1e−7 ControlledF-GLPT wire signaling by APC R-ASPT channel channel APC APC carries APCF/R 64 kb/s 2e−1 Controlled F-GLPT channel commands by APC R-ASPT

[0096] The Spreading Codes

[0097] The CDMA code generators used to encode the logical channels ofthe present invention employ Linear Shift Registers (LSRs) with feedbacklogic which is a method well known in the art. The code generators ofthe present embodiment of the invention generate 64 synchronous uniquesequences. Each RF communication channel uses a pair of these sequencesfor complex spreading (in- phase and quadrature) of the logicalchannels, so the generator gives 32 complex spreading sequences. Thesequences are generated by a single seed which is initially loaded intoa shift register circuit.

[0098] The Generation of Spreading Code Sequences and Seed Selection

[0099] The spreading code period of the present invention is defined asan integer multiple of the symbol duration, and the beginning of thecode period is also the beginning of the symbol. The relation betweenbandwidths and the symbol lengths chosen for the exemplary embodiment ofthe present invention is: BW (MHZ) L(chips/symbol) 7  91 10 130 10.5 13314 182 15 195

[0100] The spreading code length is also a multiple of 64 and of 96 forISDN frame support. The spreading code is a sequence of symbols, calledchips or chip values. The general methods of generating pseudorandomsequences using Galois Field mathematics is known to those skilled inthe art; however, a unique set, or family, of code sequences has beenderived for the present invention. First, the length of the linearfeedback shift register to generate a code sequence is chosen, and theinitial value of the register is called a “seed”. Second, the constraintis imposed that no code sequence generated by a code seed may be acyclic shift of another code sequence generated by the same code seed.Finally, no code sequence generated from one seed may be a cyclic shiftof a code sequence generated by another seed.

[0101] It has been determined that the spreading code length of chipvalues of the present invention is:

128×233415=29877120  (1)

[0102] The spreading codes are generated by combining a linear sequenceof period 233415 and a nonlinear sequence of period 128

[0103] The FBCH channel of the exemplary embodiment is an exceptionbecause it is not coded with the 128 length sequence, so the FBCHchannel spreading code has period 233415.

[0104] Producing a nonlinear sequence of length 128 may be implementedin several different ways. First, the nonlinear sequence may begenerated using a linear feedback shift register: a fixed sequenceloaded into a shift register with a feed-back connection. The fixedsequence can be generated by an m-sequence of length 127 padded with anextra logic 0, 1, or random value using clock suppression and a logiccircuit, as is well known in the art. However, generation of a sequencein real time in this manner may present timing and delay issues, as wellas increasing complexity required to provide a desired phase of thesequence.

[0105] Consequently, in the exemplary embodiment of the presentinvention, the values of the nonlinear sequence of length 128 aregenerated first and then stored in memory within the system. Thenonlinear sequence then may be provided, for example, by playing thesequence values from the memory. Another embodiment of the presentinvention includes loading the stored nonlinear-sequence into a shiftregister with feedback from the last to the first stage. The nonlinearsequence then is repetitively cycled through the shift register and anydesired phase of the nonlinear sequence may be provided from thecorresponding shift register stage.

[0106] The linear sequence of length L=233415 is generated using alinear feedback shift register (LFSR) circuit with 36 stages. Thefeedback connections correspond to a irreducible polynomial h(n) ofdegree 36. The polynomial h(x) chosen for the exemplary embodiment ofthe present invention is

h(x)=x ³⁶ +x ³⁵ +x ³⁰ +x ²⁸ +x ²⁶ +x ²⁵ +x ²² +x ²⁰ +x ¹⁹ +X ¹⁷ +x ¹⁶ +x¹⁵ +x ¹⁴ +x ¹² +x ¹¹ +x ⁹ +x ⁸ +x ⁴ +x ³ +x ²+1

[0107] or, in binary notation

h(x)=(1100001010110010110111101101100011101)  (2)

[0108] A group of “seed” values for a LFSR representing the polynomialh(x) of equation (2) which generates code sequences that are nearlyorthogonal with each other is determined. The first requirement of theseed values is that the seed values do not generate two code sequenceswhich are simply cyclic shifts of each other.

[0109] The seeds are represented as elements of GF(2³⁶) which is thefield of residue classes modulo h(x). This field has a primitive elementδ=x²+x+1. The binary representation of δ is

δ=000000000000000000000000000000000111  (3)

[0110] Every element of GF(2³⁶) can also be written as a power of δreduced modulo h(x). Consequently, the seeds are represented as powersof δ, the primitive element.

[0111] The solution for the order of an element does not require asearch of all values; the order of an element divides the order of thefield (GF(2³⁶)). When δ is any element of GF(2³⁶) with

x^(e)≡1  (4)

[0112] for some e, then e|2³⁶−1. Therefore, the order of any element inGF(2³⁶) divides 2³⁶−1.

[0113] Using these constraints, it has been determined that a numericalsearch generates a group of seed values, n, which are powers of δ, theprimitive element of h(x).

[0114] The present invention includes a method to increase the number ofavailable seeds for use in a CDMA communication system by recognizingthat certain cyclic shifts of the previously determined code sequencesmay be used simultaneously. The round trip delay for the cell sizes andbandwidths of the present invention are less than 3000 chips. In oneembodiment of the present invention, sufficiently separated cyclicshifts of a sequence can be used within the same cell without causingambiguity for a receiver attempting to determine the code sequence. Thismethod enlarges the set of sequences available for use.

[0115] By implementing the tests previously described, a total of 3879primary seeds were determined through numerical computation. These seedsare given mathematically as

δ^(n) modulo h(x)  (5)

[0116] where 3879 values of n are listed in the Appendix A, with δ=(00,. . . 00111) as before in (3).

[0117] When all primary seeds are known, all secondary seeds of thepresent invention are derived from the primary seeds by shifting themmultiples of 4095 chips modulo h(x). Once a family of seed values isdetermined, these values are stored in memory and assigned to logicalchannels as necessary. Once assigned, the initial seed value is simplyloaded into LFSR to produce the required spreading code sequenceassociated with the seed value.

[0118] Rapid Acquisition Feature of Long and Short Codes.

[0119] Rapid acquisition of the correct code phase by a spread-spectrumreceiver is improved by designing spreading codes which are faster todetect. The present embodiment of the invention includes a new method ofgenerating code sequences that have rapid acquisition properties byusing one or more of the following methods. First, a long code may beconstructed from two or more short codes. The new implementation usesmany code sequences, one or more of which are rapid acquisitionsequences of length L that have average acquisition phase searchesr=log2L. Sequences with such properties are well known to thosepracticed in the art. The average number of acquisition test phases ofthe resulting long sequence is a multiple of r=log2L rather than half ofthe number of phases of the long sequence.

[0120] Second, a method of transmitting complex valued spreading codesequences (In-phase (I) and Quadrature (Q) sequences) in a pilotspreading code signal may be used rather than transmitting real valuedsequences. Two or more separate code sequences may be transmitted overthe complex channels. If the sequences have different phases, anacquisition may be done by acquisition circuits in parallel over thedifferent code sequences when the relative phase shift between the twoor more code channels is known. For example, for two sequences, one canbe sent on an In phase (1) channel and one on the Quadrature (Q)channel. To search the code sequences, the acquisition detection meanssearches the two channels, but begins the (Q) channel with an offsetequal to one-half of the spreading code sequence length. With codesequence length of N, the acquisition means starts the search at N/2 onthe (Q) channel. The average number of tests to find acquisition is N/2for a single code search, but searching the (I) and phase delayed (Q)channel in parallel reduces the average number of tests to N/4. Thecodes sent on each channel could be the same code, the same code withone channel's code phase delayed, or different code sequences.

[0121] Epoch and Sub-epoch Structures

[0122] The long complex spreading codes used for the exemplary system ofthe present invention have a number of chips after which the coderepeats. The repetition period of the spreading sequence is called anepoch. To map the logical channels to CDMA spreading codes, the presentinvention uses an Epoch and Sub-epoch structure. The code period for theCDMA spreading code to modulate logical channels is 29877120 chips/codeperiod which is the same number of chips for all bandwidths. The codeperiod is the epoch of the present invention, and Table 3 below definesthe epoch duration for the supported chip rates. In addition, twosub-epochs are defined over the spreading code epoch and are 233415chips and 128 chips long.

[0123] The 233415 chip sub-epoch is referred to as a long sub-epoch, andis used for synchronizing events on the RF communication interface suchas encryption key switching and changing from global to assigned codes.The 128 chip short epoch is defined for use as an additional timingreference. The highest symbol rate used with a single CDMA code is 64ksym/s. There is always an integer number of chips in a symbol durationfor the supported symbol rates 64, 32, 16, and 8 ksym/s. TABLE 3Bandwidths, Chip Rates, and Epochs number of 128 chip 233415 chip ChipRate, chips in a sub-epoch sub-epoch Epoch Bandwidth Complex 64 kbit/secduration* duration* duration (MHz) (Mchip/sec) symbol (μs) (ms) (sec) 75.824 91 21.978 40.078 5.130 10 8.320 130 15.385 28.055 3.591 10.5 8.512133 15.038 27.422 3.510 14 11.648 182 10.989 20.039 2.565 15 12.480 19510.256 18.703 2.394

[0124] Mapping of the Logical Channels to Epochs and Sub-epochs

[0125] The complex spreading codes are designed such that the beginningof the sequence epoch coincides with the beginning of a symbol for allof the bandwidths supported. The present invention supports bandwidthsof 7, 10, 10.5, 14, and 15 MHz. Assuming nominal 20% roll-off, thesebandwidths correspond to the following chip rates in Table 4. TABLE 4Supported Bandwidths and Chip Rates for CDMA. BW R_(c) (ComplexFactorization (MHz) Mchips/sec) Excess BW, % L: (R_(c)/L) = 64k of L 75.824 20.19  91 7 × 13 10 8.320 20.19 130 2 × 5 × 13 10.5 8.512 23.36133 7 × 19 14 11.648 20.19 182 2 × 7 × 13 15 12.480 20.19 195 3 × 5 × 13

[0126] The number of chips in an epoch is:

N=29877120=2⁷ x3³ x5x7x13x19  (6)

[0127] If interleaving is used, the beginning of an interleaver periodcoincides with the beginning of the sequence epoch. The spreadingsequences generated using the method of the present invention cansupport interleaver periods that are multiples of 1.5 ms for variousbandwidths.

[0128] Cyclic sequences of the prior art are generated using linearfeedback shift register (LFSR) circuits. However, this method does notgenerate sequences of even length. One embodiment of the spreading codesequence generator using the code seeds generated previously is shown inFIG. 2a, FIG. 2b, and FIG. 2c. The present invention uses a 36 stageLFSR 201 to generate a sequence of period N′=233415=3³x5x7x13x19, whichis C_(o) in FIG. 2a. In FIGS. 2a, 2 b, and 2 c, the symbol ⊕ representsa binary addition (EXCLUSIVE-OR). A sequence generator designed as abovegenerates the in-phase and quadrature parts of a set of complexsequences. The tap connections and initial state of the 36 stage LFSRdetermine the sequence generated by this circuit. The tap coefficientsof the 36 stage LFSR are determined such that the resulting sequenceshave the period 233415. Note that the tap connections shown in FIG. 2acorrespond to the polynomial given in equation (2). Each resultingsequence is then overlaid by binary addition with the 128 lengthsequence C_(*) to obtain the epoch period 29877120.

[0129]FIG. 2b shows a Feed Forward (FF) circuit 202 which is used in thecode generator. The signal X[n−1] is output of the chip delay 211, andthe input of the chip delay 211 is X[n]. The code chip C[n] is formed bythe logical adder 212 from the input X[n] and X[n−1]. FIG. 2c shows thecomplete spreading code generator. From the LFSR 201, output signals gothrough a chain of up to 63 single stage FFs 203 cascaded as shown. Theoutput of each FF is overlaid with the short, even code sequence C_(*)period 128=2⁷ which is stored in code memory 222 and which exhibitsspectral characteristics of a pseudorandom sequence to obtain the epochN=29877120. This sequence of 128 is determined by using an m-sequence(PN sequence) of length 127=2⁷−1 and adding a bit-value, such as logic0, to the sequence to increase the length to 128 chips. The even codesequence C_(*) is input to the even code shift register 221, which is acyclic register, that continually outputs the sequence. The shortsequence is then combined with the long sequence using an EXCLUSIVE-ORoperation 213, 214, 220.

[0130] As shown in FIG. 2c, up to 63 spreading code sequences C₀ throughC₆₃ are generated by tapping the output signals of FFs 203 and logicallyadding the short sequence C_(*) in binary adders 213, 214, and 220, forexample. One skilled in the art would realize that the implementation ofFF 203 will create a cumulative delay effect for the code sequencesproduced at each FF stage in the chain. This delay is due to the nonzeroelectrical delay in the electronic components of the implementation. Thetiming problems associated with the delay can be mitigated by insertingadditional delay elements into the FF chain in one version of theembodiment of the invention. The FF chain of FIG. 2c with additionaldelay elements is shown in FIG. 2d.

[0131] The code-generators in the exemplary embodiment of the presentinvention are configured to generate either global codes, or assignedcodes. Global codes are CDMA codes that can be received or transmittedby all users of the system. Assigned codes are CDMA codes that areallocated for a particular connection. When a set of sequences aregenerated from the same generator as described, only the seed of the 36stage LFSR is specified to generate a family of sequences. Sequences forall the global codes, are generated using the same LFSR circuit.Therefore, once an SU has synchronized to the Global pilot signal froman RCS and knows the seed for the LFSR circuit for the Global Channelcodes, it can generate not only the pilot sequence but also all otherglobal codes used by the RCS.

[0132] The signal that is upconverted to RF is generated as follows. Theoutput signals of the above shift register circuits are converted to anantipodal sequence (0 maps into +1, 1 maps into −1). The Logicalchannels are initially converted to QPSK signals, which are mapped asconstellation points as is well known in the art. The In-phase andQuadrature channels of each QPSK signal form the real and imaginaryparts of the complex data value. Similarly, two spreading codes are usedto form complex spreading chip values. The complex data are spread bybeing multiplied by the complex spreading code. Similarly, the receivedcomplex data is correlated with the conjugate of the complex spreadingcode to recover despread data.

[0133] Short Codes

[0134] Short codes are used for the initial ramp-up process when an SUaccesses an RCS. The period of the short codes is equal to the symbolduration and the start of each period is aligned with a symbol boundary.Both SU and RCS derive the real and imaginary parts of the short codesfrom the last eight feed-forward sections of the sequence generatorproducing the global codes for that cell.

[0135] The short codes that are in use in the exemplary embodiment ofthe invention are updated every 3 ms. Other update times that areconsistent with the symbol rate may be used. Therefore, a change-overoccurs every 3 ms starting from the epoch boundary. At a change-over,the next symbol length portion of the corresponding feed-forward outputbecomes the short code. When the SU needs to use a particular shortcode, it waits until the first 3 ms boundary of the next epoch andstores the next symbol length portion output from the corresponding FFsection. This shall be used as the short code until the nextchange-over, which occurs 3 ms later.

[0136] The signals represented by these short codes are known as ShortAccess Channel pilots (SAXPTs).

[0137] Mapping of Logical Channels to Spreading Codes

[0138] The exact relationship between the spreading code sequences andthe CDMA logical channels and pilot signals is documented in Table 5aand Table 5b. Those signal names ending in ‘-CH’ correspond to logicalchannels. Those signal names ending in ‘-PT’ correspond to pilotsignals, which are described in detail below. TABLE 5A Spreading codesequences and global CDMA codes Logical Channel or Sequence QuadraturePilot Signal Direction C₀ I FBCH Forward (F) C₁ Q FBCH F C₂⊕C* I GLPT FC₃⊕C* Q GLPT F C₄⊕C* I SBCH F C₅⊕C* Q SBCH F C₆⊕C* I CTCH (0) F C₇⊕C* QCTCH (0) F C₈⊕C* I APCH (1) F C₉⊕C* Q APCH (1) F C₁₀⊕C_(*) I CTCH (1) FC₁₁⊕C_(*) Q CTCH (1) F C₁₂⊕C_(*) I APCH (1) F C₁₃⊕C_(*) Q APCH (1) FC₁₄⊕C_(*) I CTCH (2) F C₁₅⊕C_(*) Q CTCH (2) F C₁₆⊕C_(*) I APCH (2) FC₁₇⊕C_(*) Q APCH (2) F C₁₈⊕C_(*) I CTCH (3) F C₁₉⊕C_(*) Q CTCH (3) FC₂₀⊕C_(*) I APCH (3) F C₂₁⊕C_(*) Q APCH (3) F C₂₂⊕C_(*) I reserved —C₂₃⊕C_(*) Q reserved — . . . . . . . . . . . . C₄₀⊕C_(*) I reserved —C₄₁⊕C_(*) Q reserved — C₄₂⊕C_(*) I AXCH(3) Reverse (R) C₄₃⊕C_(*) QAXCH(3) R C₄₄⊕C_(*) I LAXPT(3) R SAXPT(3) seed C₄₅⊕C_(*) Q LAXPT(3) RSAXPT(3) seed C₄₆⊕C_(*) I AXCH(2) R C₄₇⊕C_(*) Q AXCH(2) R C₄₈⊕C_(*) ILAXPT(2) R SAXPT(2) seed C₄₉⊕C_(*) Q LAXPT(2) R SAXPT(2) seed C₅₀⊕C_(*)I AXCH(1) R C₅₁⊕C_(*) Q AXCH(1) R C₅₂⊕C_(*) I LAXPT(1) R SAXPT(1) seedC₅₃⊕C_(*) Q LAXPT(1) R SAXPT(1) seed C₅₄⊕C_(*) I AXCH(0) R C₅₅⊕C_(*) QAXCH(0) R C₅₆⊕C_(*) I LAXPT(0) R SAXPT(0) seed C₅₇⊕C_(*) Q LAXPT(0) RSAXPT(0) seed C58⊕C_(*) I IDLE — C59⊕C_(*) Q IDLE — C60⊕C_(*) I AUX RC61⊕C_(*) Q AUX R C62⊕C_(*) I reserved — C63⊕C_(*) Q reserved —

[0139] TABLE 5B Spreading code sequences and assigned CDMA codes.Logical Channel or Sequence Quadrature Pilot Signal Direction C₀⊕C_(*) IASPT Reverse (R) C₁⊕C_(*) Q ASPT R C₂⊕C_(*) I APCH R C₃⊕C_(*) Q APCH RC₄⊕C_(*) I OWCH R C₅⊕C_(*) Q OWCH R C₆⊕C_(*) I TRCH(0) R C₇⊕C_(*) QTRCH(0) R C₈⊕C_(*) I TRCH(1) R C₉⊕C Q TRCH(1) R C₁₀⊕C_(*) I TRCH(2) RC₁₁⊕C_(*) Q TRCH(2) R C₁₂⊕C_(*) I TRCH(3) R C₁₃⊕C_(*) Q TRCH(3) RC₁₄⊕C_(*) I reserved — C₁₅⊕C_(*) Q reserved — . . . . . . . . . . . .C₄₄⊕C_(*) I reserved — C₄₅⊕C_(*) Q reserved — C₄₆⊕C_(*) I TRCH(3)Forward (F) C₄₇⊕C_(*) Q TRCH(3) F C₄₈⊕C_(*) I TRCH(2) F C₄₉⊕C_(*) QTRCH(2) F C₅₀⊕C_(*) I TRCH(1) F C₅₁⊕C_(*) Q TRCH(1) F C₅₂⊕C_(*) ITRCH(0) F C₅₃⊕C_(*) Q TRCH(0) F C₅₄⊕C_(*) I OWCH F C₅₅⊕C_(*) Q OWCH FC₅₆⊕C_(*) I APCH F C₅₇⊕C_(*) Q APCH F C₅₈⊕C_(*) I IDLE — C₅₉⊕C_(*) QIDLE — C₆₀⊕C_(*) I reserved — C₆₁⊕C_(*) Q reserved — C₆₂⊕C_(*) Ireserved — C₆₃⊕C_(*) Q reserved —

[0140] For global codes, the seed values for the 36 bit shift registerare chosen to avoid using the same code, or any cyclic shift of the samecode, within the same geographical area to prevent ambiguity or harmfulinterference. No assigned code is equal to, or a cyclic shift of aglobal code.

[0141] Pilot Signals

[0142] The pilot signals are used for synchronization, carrier phaserecovery, and for estimating the impulse response of the radio channel.

[0143] The RCS 104 transmits a forward link pilot carrier reference as acomplex pilot code sequence to provide time and phase reference for allSUs 111, 112, 115, 117 and 118 in its service area. The power level ofthe Global Pilot (GLPT) signal is set to provide adequate coverage overthe whole RCS service area, which area depends on the cell size. Withonly one pilot signal in the forward link, the reduction in systemcapacity due to the pilot energy is negligible.

[0144] The SUs 111, 112, 115, 117 and 118 each transmits a pilot carrierreference as a quadrature modulated (complex-valued) pilot spreadingcode sequence to provide a time and phase reference to the RCS for thereverse link. The pilot signal transmitted by the SU of one embodimentof the invention is 6 dB lower than the power of the 32 kb/s POTStraffic channel. The reverse pilot channel is subject to APC. Thereverse link pilot associated with a particular connection is called theAssigned Pilot (ASPT). In addition, there are pilot signals associatedwith access channels. These are called the Long Access Channel Pilots(LAXPTs). Short access channel pilots (SAXPTs) are also associated withthe access channels and used for spreading code acquisition and initialpower ramp-up

[0145] All pilot signals are formed from complex codes, as definedbelow:

GLPT (forward)={C ₂ ⊕C _(*))+j.(C ₃ ⊕C _(*))}.{(1)+j.(0)}

{Complex Code}.{Carrier}

[0146] The complex pilot signals are de-spread by multiplication withconjugate spreading codes: {(C₂⊕C_(*))−j.(C₃⊕C_(*))}. By contrast,traffic channels are of the form:

TRCH _(n) (forward/reverse)={(C _(k) ⊕C _(*))+j.(C ₁ ⊕C_(*))}.{(±1)+j(±1)}

{Complex Codes}.{Data Symbol}

[0147] which thus form a constellation set at $\frac{\pi}{4}$

[0148] radians with respect to the pilot signal constellations.

[0149] The GLPT constellation is shown in FIG. 3a, and the TRCH_(n)traffic channel constellation is shown in FIG. 3b.

[0150] Logical Channel Assignment of the FBCH, SBCH, and TrafficChannels

[0151] The FBCH is a global forward link channel used to broadcastdynamic information about the availability of services and AXCHs.Messages are sent continuously over this channel, and each message lastsapproximately 1 ms. The FBCH message is 16 bits long, repeatedcontinuously, and is epoch aligned. The FBCH is formatted as defined inTable 6. TABLE 6 FBCH format Bit Definition 0 Traffic Light 0 1 TrafficLight 1 2 Traffic Light 2 3 Traffic Light 3 4-7 service indicator bits 8Traffic Light 0 9 Traffic Light 1 10  Traffic Light 2 11  Traffic Light3 12-15 service indicator bits

[0152] For the FBCH, bit 0 is transmitted first. As used in Table 6, atraffic light corresponds to an Access Channel (AXCH) and indicateswhether the particular access channel is currently in use (a red) or notin use (a green). A logic ‘1’ indicates that the traffic light is green,and a logic ‘0’ indicates the traffic light is red. The values of thetraffic light bits may change from octet to octet, and each 16 bitmessage contains distinct service indicator bits which describe thetypes of services that are available for the AXCHs.

[0153] One embodiment of the present invention uses service indicatorbits as follows to indicate the availability of services or AXCHs, Theservice indicator bits {4, 5, 6, 7, 12, 13, 14, 15} taken together maybe an unsigned binary number, with bit 4 as the MSB and bit 15 as theLSB. Each service type increment has an associated nominal measure ofthe capacity required, and the FBCH continuously broadcasts theavailable capacity. This is scaled to have a maximum value equivalent tothe largest single service increment possible. When an SU requires a newservice or an increase in the number of bearers, it compares thecapacity required to that indicated by the FBCH, and then considersitself blocked if the capacity is not available. The FBCH and thetraffic channels are aligned to the epoch.

[0154] Slow Broadcast Information frames contain system or other generalinformation that is available to all SUs and Paging Information framescontain information about call requests for particular SUs. SlowBroadcast Information frames and Paging Information frames aremultiplexed together on a single logical channel which forms the SlowBroadcast Channel (SBCH). As previously defined, the code epoch is asequence of 29 877 20 chips having an epoch duration which is a functionof the chip rate defined in Table 7 below. In order to facilitate powersaving, the channel is divided into N “Sleep” Cycles, and each Cycle issubdivided into M Slots, which are 19 ms long, except for 10.5 Mhzbandwidth which has slots of 18 ms. TABLE 7 SBCH Channel Format OutlineSpreading Epoch Cycle Slots/ Slot Bandwidth Code Rate Length Cycles/Length Cycle Length (MHz) (MHz) (ms) Epoch N (ms) M (ms) 7.0 5.824 51305 1026 54 19 10.0 8.320 3591 3 1197 63 19 10.5 8.512 3510 3 1170 65 1814.0 11.648 2565 3 855 45 19 15.0 12.480 2394 2 1197 63 19

[0155] Sleep Cycle Slot #1 is always used for slow broadcastinformation. Slots #2 to #M−1 are used for paging groups unless extendedslow broadcast information is inserted. The pattern of cycles and slotsin one embodiment of the present invention run continuously at 16 kb/s.

[0156] Within each Sleep Cycle the SU powers-up the receiver andre-acquires the pilot code. It then achieves carrier lock to asufficient precision for satisfactory demodulation and Viterbi decoding.The settling time to achieve carrier lock may be up to 3 Slots induration. For example, an SU assigned to Slot #7 powers up the Receiverat the start of Slot #4. Having monitored its Slot the SU will haveeither recognized its Paging Address and initiated an access request, orfailed to recognize its Paging Address in which case it reverts to theSleep mode. Table 8 shows duty cycles for the different bandwidths,assuming a wake-up duration of 3 Slots. TABLE 8 Sleep-Cycle Power SavingBandwidth (MHz) Slots/Cycle Duty Cycle  7.0 54 7.4% 10.0 63 6.3% 10.5 656.2% 14.0 45 8.9% 15.0 63 6.3%

Spreading Code Tracking and AMF Detection in Multipath Channels

[0157] Spreading Code Tracking

[0158] Three CDMA spreading code tracking methods in multipath fadingenvironments are described which track the code phase of a receivedmultipath spread-spectrum signal. The first is the prior art trackingcircuit which simply tracks the spreading code phase with the highestdetector output signal value, the second is a tracking circuit thattracks the median value of the code phase of the group of multipathsignals, and the third is the centroid tracking circuit which tracks thecode-phase of an optimized, least mean squared weighted average of themultipath signal components. The following describes the algorithms bywhich the spreading code phase of the received CDMA signal is tracked.

[0159] A tracking circuit has operating characteristics that reveal therelationship between the time error and the control voltage that drivesa Voltage Controlled Oscillator (VCO) of a spreading code phase trackingcircuit. When there is a positive timing error, the tracking circuitgenerates a negative control voltage to offset the timing error. Whenthere is a negative timing error; the tracking circuit generates apositive control voltage to offset the timing error. When the trackingcircuit generates a zero value, this value corresponds to the perfecttime alignment called the ‘lock-point’. FIGS. 3c and 3 d show the basictracking circuit. Received signal r(t) is applied to the chip MatchedFilter 301, which maximizes the chip signal to noise ratio.

[0160] In FIG. 3c, the output signal of the chip matched filter x(t) issampled by the sampler 302 at a sampling rate of twice the chip rate toproduce samples x[nT] and x[nT+T/2]. The samples x[nT] and x[nT+T/2] areused by a tracking circuit 304 to determine if the phase of thespreading code c(t) of the code generator 303 is correct. The trackingcircuit 304 produces an error signal e(t) as an input to the codegenerator 303. The code generator 303 uses this signal e(t) as an inputsignal to adjust the code-phase it generates.

[0161]FIG. 3d shows a spreading code phase tracking system similar tothat shown in FIG. 3c, but the output signal of the chip matched filterx(t) is sampled by the sampler 306 at a sampling rate equivalent to thechip rate to produce samples x′[nT] only. The tracking circuit 308 usesthe samples x′[nT] in a manner similar to that of the tracking circuit304 of FIG. 3c. The configuration of FIG. 3d may be used to track thecode phase once an initial acquisition of the spreading code phase hasoccurred. In such a situation, the approximate chip timing can berecovered by a coarse timing recovery circuit 310 from an acquisitioncode generator clock CLK, for example, and the timing signal can be usedby the Code Generator 303 and Sampler 306 to sample the signal x(t) atthe approximate desired sampling time during a chip period.Consequently, to relate the operation of Track circuit 308 of theconfiguration of FIG. 3d to the following description of trackingmethods assuming a configuration as described in 3 c, the early samplesx[nT] when sampling at twice the chip rate become the even samples ofx′[nT] when sampling at the chip rate, and the late samples x[nT+T/2]become the odd samples of x′[nT].

[0162] In a CDMA system, the signal transmitted by the reference user iswritten in the low-pass representation as $\begin{matrix}{{s(t)} = {\sum\limits_{k = {- \infty}}^{\infty}{c_{k}{P_{Tc}\left( {t - {kT}_{c}} \right)}}}} & (7)\end{matrix}$

[0163] where c_(k) represents the spreading code coefficients, P_(Tc)(t)represents the spreading code chip waveform, and T_(c) is the chipduration. Assuming that the reference user is not transmitting data sothat only the spreading code modulates the carrier. Referring to FIG.3c, the received signal is $\begin{matrix}{{r(t)} = {\sum\limits_{i = 1}^{M}{a_{i}{s\left( {t - \tau_{i}} \right)}}}} & (8)\end{matrix}$

[0164] Here, a_(i) is due to fading effect of the multipath channel onthe i-th path and τ_(i) is the random time delay associated with thesame path. The receiver passes the received signal through a matchedfilter, which is implemented as a correlation receiver and is describedbelow. This operation is done in two steps: first the signal is passedthrough a chip matched filter and sampled to recover the spreading codechip values, then this chip sequence is correlated with the locallygenerated code sequence.

[0165]FIG. 3c shows the chip matched filter 301, matched to the chipwaveform P_(Tc)(t), and the sampler 302. Ideally, the signal x(t) at theoutput terminal of the chip matched filter is $\begin{matrix}{{x(t)} = {\sum\limits_{i = k}^{M}{\sum\limits_{k = {- \infty}}^{\infty}{a_{i}c_{k}{g\left( {t - \tau_{i} - {kT}_{c}} \right)}}}}} & (9)\end{matrix}$

[0166] where

g(t)=P _(Tc)(t)*h _(R)(t)  (10)

[0167] Here, h_(R)(t) is the impulse response of the chip matched filterand ‘*’ denotes convolution. The order of the summations can berewritten as $\begin{matrix}{{{x(t)} = {\sum\limits_{k = {- \infty}}^{\infty}{c_{k}{f\left( {t - {kT}_{c}} \right)}}}}{where}} & (11) \\{{f(t)} = {\sum\limits_{i = 1}^{M}{a_{i}{g\left( {t - \tau_{i}} \right)}}}} & (12)\end{matrix}$

[0168] In the multipath channel described above, the sampler samples theoutput signal of the matched filter to produce x(nT) at the maximumpower level points of g(t). In practice, however, the waveform g(t) isseverely distorted because of the effect of the multipath signalreception, and a perfect time alignment of the signals is not available.

[0169] When the multipath distortion in the channel is negligible and aperfect estimate of the timing is available, i.e., a=1, τt=0, anda_(i)=0, i=2, . . . , M, the received signal is r(t)=s(t). Then, withthis ideal channel model, the output of the chip matched filter becomes$\begin{matrix}{{x(t)} = {\sum\limits_{k = {- \infty}}^{\infty}{c_{k}{g\left( {t - {kT}_{c}} \right)}}}} & (13)\end{matrix}$

[0170] When there is multipath fading, however, the received spreadingcode chip value waveform is distorted, and has a number of local maximathat can change from one sampling interval to another depending on thechannel characteristics.

[0171] For multipath fading channels with quickly changing channelcharacteristics, it is not practical to try to locate the maximum of thewaveform f(t) in every chip period interval. Instead, a time referencemay be obtained from the characteristics of f(t) that may not change asquickly. Three tracking methods are described based on differentcharacteristics of f(t).

[0172] Prior Art Spreading Code Tracking Method:

[0173] Prior art tracking methods include a code tracking circuit inwhich the receiver attempts to determine the timing of the maximummatched filter output value of the chip waveform occurs and sample thesignal accordingly. However, in multipath fading channels, the receiverdespread code waveform can have a number of local maxima, especially ina mobile environment. In the following, f(t) represents the receivedsignal waveform of the spreading code chip convolved with the channelimpulse response. The frequency response characteristic of f(t) and themaximum of this characteristic can change rather quickly making itimpractical to track the maximum of f(t).

[0174] Define τ to be the time estimate that the tracking circuitcalculates during a particular sampling interval. Also, define thefollowing error function $\begin{matrix}\begin{matrix}{ɛ = \left\{ {\underset{\{{t:{|{\tau - t}|{> \delta}}}\}}{\int{{f(t)}{t}}},} \right.} & \left| {\tau - t} \middle| {> \delta} \right. \\{ɛ = 0} & \left| {\tau - t} \middle| {< \delta} \right.\end{matrix} & (14)\end{matrix}$

[0175] The tracking circuits of the prior art calculate a value of theinput signal that minimizes the error ε. One can write $\begin{matrix}{{\min \quad ɛ} = {1 - {\max\limits_{\tau}{\int_{\tau - \delta}^{\tau + \delta}{{f(t)}{t}}}}}} & (15)\end{matrix}$

[0176] Assuming f(τ) has a smooth shape in the values given, the valueof τ for which f(τ) is maximum minimizes the error ε, so the trackingcircuit tracks the maximum point of f(t).

[0177] Median Weighted Value Tracking Method:

[0178] The Median Weighted Tracking Method of one embodiment of thepresent invention, minimizes the absolute weighted error, defined as

ε=∫_(−∞) ^(∞) |t−τ|f(t)dt  (16)

[0179] This tracking method calculates the ‘median’ signal value of f(t)by collecting information from all paths, where f(τ) is as in equation12. In a multipath fading environment, the waveform f(t) can havemultiple local maxima, but only one median.

[0180] To minimize ε, take the derivative of equation (16) is taken withrespect to τ and the result is equated to zero, which gives

∫_(−∞) ^(τ) f(t)dt=∫ _(τ) ^(∞) f(t)dt  (17)

[0181] The value of τ that satisfies (17) is called the ‘median’ off(t). Therefore, the Median Tracking Method of the present embodimenttracks the median of f(t). FIG. 4 shows an implementation of thetracking circuit based on minimizing the absolute weighted error definedabove. The signal x(t) and its one-half chip offset version x(t+T/2) aresampled by the A/D 401 at a rate 1/T. The following equation determinesthe operating characteristic of the circuit in FIG. 4: $\begin{matrix}{{ɛ(\tau)} = {\sum\limits_{n = 1}^{2L}\left| {f\left( {\tau - {{nT}/2}} \right)} \middle| {- \left| {f\left( {\tau + {{nT}/2}} \right)} \right|} \right.}} & (18)\end{matrix}$

[0182] Tracking the median of a group of multipath signals keeps thereceived energy of the multipath signal components substantially equalon the early and late sides of the median point of the correct locallygenerated spreading code phase c_(n). The tracking circuit consists ofan A/D 401 which samples an input signal x(t) to form the half-chipoffset samples. The half chip offset samples are alternatively groupedinto even samples called an early set of samples x(nT+x) and odd samplescalled a late set of samples x(nT+(T/2)+τ). The first correlation bankadaptive matched filter 402 multiplies each early sample by thespreading code phases c(n+1), c(n+2), . . . , c(n+L), where L is smallcompared to the code length and approximately equal to number of chipsof delay between the earliest and latest multipath signal. The output ofeach correlator is applied to a respective first sum-and-dump bank 404.The magnitudes of the output values of the L sum-and-dumps arecalculated in the calculator 406 and then summed in summer 408 to givean output value proportional to the signal energy in the early multipathsignals. Similarly, a second correlation bank adaptive matched filter403 operates on the late samples, using code phases c(n−1), c(n−2), . .. , c(n−L), and each output signal is applied to a respectivesum-and-dump circuit in an integrator 405. The magnitudes of the Lsum-and-dump output signals are calculated in calculator 407 and thensummed in summer 409 to give a value for the late multipath signalenergy. Finally, the subtractor 410 calculates the difference andproduces error signal ε(t) of the early and late signal energy values.

[0183] The tracking circuit adjusts by means of error signal ε(τ) thelocally generated code phases c(t) to cause the difference between theearly and late values to tend toward 0.

[0184] Centroid Tracking Method

[0185] The optimal spreading code tracking circuit of one embodiment ofthe present invention is called the squared weighted tracking (orcentroid) circuit. Defining τ to denote the time estimate that thetracking circuit calculates, based on some characteristic of f(t), thecentroid tracking circuit minimizes the squared weighted error definedas

ε=∫_(−∞) ^(∞) |t−τ| ² f(t)dt  (19)

[0186] This function inside the integral has a quadratic form, which hasa unique minimum. The value of τ that minimizes ε can be found by takingthe derivative of the above equation with respect to τ and equating tozero, which gives

∫_(−∞) ^(∞)(−2t+2τ)f(t)dt=0  (20)

[0187] Therefore, the value of τ that satisfies equation (21)$\begin{matrix}{{\tau - {\frac{1}{\beta}{\int_{- \infty}^{\infty}{{{tf}(t)}{t}}}}} = 0} & (21)\end{matrix}$

[0188] is the timing estimate that the tracking circuit calculates,where β is a constant value.

[0189] Based on these observations, a realization of an exemplarytracking circuit which minimizes the squared weighted error is shown inFIG. 5a. The following equation determines the error signal ε(τ) of thecentroid tracking circuit: $\begin{matrix}{{ɛ(\tau)} = {{\sum\limits_{n = 1}^{2L}{n\left\lbrack \left| {f\left( {\tau - {{nT}/2}} \right)} \middle| {- \left| {f\left( {\tau + {{nT}/2}} \right)} \right|} \right. \right\rbrack}} = 0}} & (22)\end{matrix}$

[0190] The value that satisfies ε(τ)=0 is the perfect estimate of thetiming.

[0191] The early and late multipath signal energy on each side of thecentroid point are equal. The centroid tracking circuit shown in FIG. 5aconsists of an A/D converter 501 which samples an input signal x(t) toform the half-chip offset samples. The half chip offset samples arealternatively grouped as an early set of samples x(nT+τ) and a late setof samples x(nT+(T/2)+τ). The first correlation bank adaptive matchedfilter 502 multiplies each early sample and each late sample by thepositive spreading code phases c(n+1), c(n+2), . . . , c(n+L), where Lis small compared to the code length and approximately equal to numberof chips of delay between the earliest and latest multipath signal. Theoutput signal of each correlator is applied to a respective one of Lsum-and-dump circuits of the first sum and dump bank 504. The magnitudevalue of each sum-and-dump circuit of the sum and dump bank 504 iscalculated by the respective calculator in the calculator bank 506 andapplied to a corresponding weighting amplifier of the first weightingbank 508. The output signal of each weighting amplifier represents theweighted signal energy in a multipath component signal.

[0192] The weighted early multipath signal energy values are summed insample adder 510 to give an output value proportional to the signalenergy in the group of multipath signals corresponding to positive codephases which are the early multipath signals. Similarly, a secondcorrelation bank adaptive matched filter 503 operates on the early andlate samples, using the negative spreading code phases c(n−1), c(n−2), .. . , c(n−L); each output signal is provided to a respectivesum-and-dump circuit of discrete integrator 505. The magnitude value ofthe L sum-and-dump output signals are calculated by the respectivecalculator of calculator bank 507 and then weighted in weighting bank509. The weighted late multipath signal energy values are summed insample adder 511 to give an energy value for the group of multipathsignals corresponding to the negative code phases which are the latemultipath signals. Finally, the adder 512 calculates the difference ofthe early and late signal energy values to produce error sample value ετ).

[0193] The tracking circuit of FIG. 5a produces error signal ε(τ) whichis used to adjust the locally generated code phase c(nT) to keep theweighted average energy in the early and late multipath signal groupsequal. The embodiment shown uses weighting values that increase as thedistance from the centroid increases. The signal energy in the earliestand latest multipath signals is probably less than the multipath signalvalues near the centroid. Consequently, the difference calculated by theadder 510 is more sensitive to variations in delay of the earliest andlatest multipath signals.

[0194] Quadratic Detector for Tracking

[0195] In the new embodiment of the tracking method, the trackingcircuit adjusts sampling phase to be “optimal” and robust to multipath.Let f(t) represent the received signal waveform as in equation 12 above.The particular method of optimizing starts with a delay locked loop withan error signal ε(τ) that drives the loop. The function ε(τ) must haveonly one zero at τ=τ₀ where τ₀ is optimal. The optimal form for ε(τ) hasthe canonical form: $\begin{matrix}{{ɛ(\tau)} = {\int_{- \infty}^{\infty}{{w\left( {t,\tau} \right)}{{f(t)}}^{2}{t}}}} & (23)\end{matrix}$

[0196] where w(t, τ) is a weighting function relating f(t) to the errorε(τ), and the relationship indicated by equation (24) also holds$\begin{matrix}{{ɛ\left( {\tau + \tau_{0}} \right)} = {\int_{- \infty}^{\infty}{{w\left( {t,{\tau + \tau_{0}}} \right)}{{f(t)}}^{2}{t}}}} & (24)\end{matrix}$

[0197] It follows from equation (24) that w(t, τ) is equivalent tow(t−τ). Considering the slope M of the error signal in the neighborhoodof a lock point τ₀: $\begin{matrix}{{{M = \frac{{ɛ(\tau)}}{\tau}}}_{r_{0}} = {- {\int_{- \infty}^{\infty}{{w^{\prime}\left( {t - \tau_{0}} \right)}{g(t)}{t}}}}} & (25)\end{matrix}$

[0198] where w′(t, τ) is the derivative of w(t, τ) with respect to t,and g(t) is the average of |f(t)|².

[0199] The error ε(τ) has a deterministic part and a noise part. Let zdenote the noise component in ε(τ), then |z|² is the average noise powerin the error function ε(τ). Consequently, the optimal tracking circuitmaximizes the ratio $\begin{matrix}{F = \frac{M^{2}}{|z|^{2}}} & (26)\end{matrix}$

[0200] The implementation of the Quadratic Detector is now described.The discrete error value e of an error signal ε(τ) is generated byperforming the operation

e=y^(T)By  (27)

[0201] where the vector y represents the received signal components yi,i=0, 1, . . . L−1, as shown in FIG. 5b. The matrix B is an L by L matrixand the elements are determined by calculating values such that theratio F of equation (26) is maximized.

[0202] The Quadratic Detector described above may be used to implementthe centroid tracking system described above with reference to FIG. 5a.For this implementation, the vector y is the output signal of the sumand dump circuits 504: y={f(τ−LT), f(τ−LT+T/2), f(τ−(L−1)T), • • • f(τ),f(τ+T/2), f(τ+T), • • • f(τ+LT)} and the matrix B is set forth in table9. TABLE 9 B matrix for quadratic form of Centroid Tracking System L 0 00 0 0 0 0 0 0 0 0 L − 1/2 0 0 0 0 0 0 0 0 0 0 0 L − 1 0 0 0 0 0 0 0 0 .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 0 0 01/2 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 −1/2 0 0 0 0 . . . . .. . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 0 0 0 0 0 0 0−L + 1 0 0 0 0 0 0 0 0 0 0 0 −L + 0 1/2 0 0 0 0 0 0 0 0 0 0 −L

[0203] Determining the Minimum Value of L Needed:

[0204] The value of L in the previous section determines the minimumnumber of correlators and sum-and-dump elements. L is chosen as small aspossible without compromising the functionality of the tracking circuit.

[0205] The multipath characteristic of the channel is such that thereceived chip waveform f(t) is spread over QT_(c) seconds, or themultipath components occupy a time period of Q chips duration. The valueof L chosen is L=Q. Q is found by measuring the particular RF channeltransmission characteristics to determine the earliest and latestmultipath component signal propagation delay. QT_(c) is the differencebetween the earliest and latest multipath component arrival time at areceiver.

[0206] Adaptive Vector Correlator

[0207] An embodiment of the present invention uses an adaptive vectorcorrelator (AVC) to estimate the channel impulse response and to obtaina reference value for coherent combining of received multipath signalcomponents. The described embodiment employs an array of correlators toestimate the complex channel response affecting each multipathcomponent. The receiver compensates for the channel response andcoherently combines the received multipath signal components. Thisapproach is referred to as maximal ratio combining.

[0208] Referring to FIG. 6, the input signal x(t) to the system includesinterference noise of other message channels, multipath signals of themessage channels, thermal noise, and multipath signals of the pilotsignal. The signal is provided to AVC 601 which, in the exemplaryembodiment, includes a despreading means 602, channel estimation meansfor estimating the channel response 604, correction means for correctinga signal for effects of the channel response 603, and adder 605. The AVCdespreading means 602 is composed of multiple code correlators, witheach correlator using a different phase of the pilot code c(t) providedby the pilot code generator 608. The output signal of this despreadingmeans corresponds to a noise power level if the local pilot code of thedespreading means is not in phase with the input code signal.Alternatively, it corresponds to a received pilot signal power levelplus noise power level if the phases of the input pilot code and locallygenerated pilot code are the same. In one embodiment, as shown in FIG.6, the output signals of the correlators of the despreading means arecorrected for the channel response by the correction means 603 and areapplied to the adder 605 which collects all multipath pilot signalpower. In another embodiment, not shown, the signal x(t) and not thedespread signal is provided directly to the correction means 603, and isthen despread by a despreading circuit similar to despreading means 602.

[0209] The channel response estimation means 604 receives the combinedpilot signal and the output signals of the despreading means 602, andprovides a channel response estimate signal, w(t), to the correctionmeans 603 of the AVC, and the estimate signal w(t) is also available tothe adaptive matched filter (AMF) described below. The output signal ofthe despreading means 602 is also provided to the acquisition decisionmeans 606 which decides, based on a particular algorithm such as asequential probability ratio test (SPRT), if the present output levelsof the despreading circuits correspond to synchronization of the locallygenerated code to the desired input code phase. If the detector finds nosynchronization, then the acquisition decision means sends a controlsignal a(t) to the local pilot code generator 608 to offset its phase byone or more chip periods. When synchronization is found, the acquisitiondecision means informs tracking circuit 607, which achieves andmaintains a close synchronization between the received and locallygenerated code sequences.

[0210] An exemplary implementation of the Pilot AVC used to despread thepilot spreading code is shown in FIG. 7. The described embodimentassumes that the input signal x(t) has been sampled with sampling periodT to form samples x(nT+τ), and is composed of interference noise ofother message channels, multipath signals of message channels, thermalnoise, and multipath signals of the pilot code. The signal x(nT+τ) isapplied to L correlators, where L is the number of code phases overwhich the uncertainty within the multipath signals exists. Eachcorrelator 701, 702, 703 comprises a multiplier 704, 705, 706, whichmultiples the input signal with a particular phase of the Pilotspreading code signal c((n+i)T), and sum-and-dump circuits 708, 709,710. The output signal of each multiplier 704, 705, 706 is applied to arespective sum-and dump circuit 708, 709, 710 to perform discreteintegration. Before summing the signal energy contained in the outputsof the correlators, the AVC compensates for the channel response and thecarrier phase rotation of the different multipath signals. Each outputof each sum-and-dump 708, 709, 710 is multiplied with a derotationphaser [complex conjugate of ep(nT)] from digital phase lock loop (DPLL)721 by the respective multiplier 714, 715, 716 to account for the phaseand frequency offset of the carrier signal. The Pilot Rake AMFcalculates the weighting factors wk, k=1, . . . , L, for each multipathsignal by passing the output of each multiplier 714, 715, 716 through alow pass filter (LPF) 711, 712, 713. Each despread multipath signal ismultiplied by its corresponding weighting factor in a respectivemultiplier 717, 718, 719. The output signals of the multipliers 717,718, 719 are summed in a master adder 720, and the output signal p(nT)of the accumulator 720 consists of the combined despread multipath pilotsignals in noise. The output signal p(nT) is also input to the DPLL 721to produce the error signal ep(nT) for tracking of the carrier phase.

[0211]FIGS. 8a and 8 b show alternate embodiments of the AVC which canbe used for detection and multipath signal component combining. Themessage signal AVCs of FIGS. 8a and 8 b use the weighting factorsproduced by the Pilot AVC to correct the message data multipath signals.The spreading code signal, c(nT) is the spreading code spreadingsequence used by a particular message channel and is synchronous withthe pilot spreading code signal. The value L is the number ofcorrelators in the AVC circuit.

[0212] The circuit of FIG. 8a calculates the decision variable Z whichis given by $\begin{matrix}{Z = {{w_{1}{\sum\limits_{i = 1}^{N}{{x\left( {{iT} + \tau} \right)}{c({iT})}}}} + {w_{2}{\sum\limits_{i = 1}^{N}{{x\left( {{iT} + \tau} \right)}{c\left( {\left( {i + 1} \right)T} \right)}}}} + \ldots + {w_{L}{\sum\limits_{i = 1}^{L}{x\left( {{iT} + \tau} \right)}}} + {c\left( {\left( {i + L} \right)T} \right)}}} & (28)\end{matrix}$

[0213] where N is the number of chips in the correlation window.Equivalently, the decision statistic is given by $\begin{matrix}{Z = {{{{x\left( {T + \tau} \right)}{\sum\limits_{i = 1}^{L}{w_{1}{c({iT})}}}} + {{x\left( {{2T} + \tau} \right)}{\sum\limits_{i = 1}^{L}{w_{1}{c\left( {\left( {i + 1} \right)T} \right)}}}} + \ldots + {{x\left( {{NT} + \tau} \right)}{\sum\limits_{i = 1}^{L}{w_{N}{c\left( {\left( {i + N} \right)T} \right)}}}}} = {\sum\limits_{k = 1}^{N}{{x\left( {{kT} - \tau} \right)}{\sum\limits_{i = 1}^{L}{w_{k}{c\left( {\left( {i + k - 1} \right)T} \right)}}}}}}} & (29)\end{matrix}$

[0214] The alternative implementation that results from equation (29) isshown in FIG. 8b.

[0215] Referring to FIG. 8a, the input signal x(t) is sampled to formx(nT+τ), and is composed of interference noise of other messagechannels, multipath signals of message channels, thermal noise, andmultipath signals of the pilot code. The signal x(nT+τ) is applied to Lcorrelators, where L is the number of code phases over which theuncertainty within the multipath signals exists. Each correlator 801,802, 803 comprises a multiplier 804, 805, 806, which multiples the inputsignal by a particular phase of the message channel spreading codesignal, and a respective sum-and-dump circuit 808, 809, 810. The outputsignal of each multiplier 804, 805, 806 is applied to a respectivesum-and dump circuit 808, 809, 810 which performs discrete integration.Before summing the signal energy contained in the output signals of thecorrelators, the AVC compensates for the different multipath signals.Each despread multipath signal and its corresponding weighting factor,which is obtained from the corresponding multipath weighting factor ofthe pilot AVC, are multiplied in a respective multiplier 817, 818, 819.The output signals of multipliers 817, 818, 819 are summed in a masteradder 820, and the output signal z(nT) of the accumulator 820 consistsof sampled levels of a despread message signal in noise.

[0216] The alternative embodiment of the invention includes a newimplementation of the AVC despreading circuit for the message channelswhich performs the sum-and-dump for each multipath signal componentsimultaneously. The advantage of this circuit is that only one sum-anddump circuit and one adder is necessary. Referring to FIG. 8b, themessage code sequence generator 830 provides a message code sequence toshift register 831 of length L. The output signal of each register 832,833, 834, 835 of the shift register 831 corresponds to the message codesequence shifted in phase by one chip. The output value of each register832, 833, 834, 835 is multiplied in multipliers 836, 837, 838, 839 withthe corresponding weighting factor w_(k), k=1, . . . , L obtained fromthe Pilot AVC. The output signals of the L multipliers 836, 837, 838,839 are summed by the adding circuit 840. The adding circuit outputsignal and the receiver input signal x(nT+τ) are then multiplied in themultiplier 841 and integrated by the sum-and-dump circuit 842 to producemessage signal z(nT).

[0217] A third embodiment of the adaptive vector correlator is shown inFIG. 8c. The embodiment shown uses the least mean square (LMS) statisticto implement the vector correlator and determines the derotation factorsfor each multipath component from the received multipath signal. The AVCof FIG. 8c is similar to the exemplary implementation of the Pilot AVCused to despread the pilot spreading code shown in FIG. 7. The digitalphase locked loop 721 is replaced by the phase locked loop 850 havingvoltage controlled oscillator 851, loop filter 852, limiter 853, andimaginary component separator 854. The difference between the correcteddespread output signal dos and an ideal despread output signal isprovided by adder 855, and the difference signal is a despread errorvalue ide which is further used by the derotation circuits to compensatefor errors in the derotation factors.

[0218] In a multipath signal environment, the signal energy of atransmitted symbol is spread out over the multipath signal components.The advantage of multipath signal addition is that a substantial portionof signal energy is recovered in an output signal from the AVC.Consequently, a detection circuit has an input signal from the AVC witha higher signal-to-noise ratio (SNR), and so can detect the presence ofa symbol with a lower bit-error ratio (BER). In addition, measuring theoutput of the AVC is a good indication of the transmit power of thetransmitter, and a good measure of the system's interference noise.

[0219] Adaptive Matched Filter

[0220] One embodiment of the current invention includes an AdaptiveMatched Filter (AMF) to optimally combine the multipath signalcomponents in a received spread spectrum message signal. The AMF is atapped delay line which holds shifted values of the sampled messagesignal and combines these after correcting for the channel response. Thecorrection for the channel response is done using the channel responseestimate calculated in the AVC which operates on the Pilot sequencesignal. The output signal of the AMF is the combination of the multipathcomponents which are summed to give a maximum value. This combinationcorrects for the distortion of multipath signal reception. The variousmessage despreading circuits operate on this combined multipathcomponent signal from the AMF.

[0221]FIG. 8d shows an exemplary embodiment of the AMF. The sampledsignal from the A/D converter 870 is applied to the L-stage delay line872. Each stage of this delay line 872 holds the signal corresponding toa different multipath signal component. Correction for the channelresponse is applied to each delayed signal component by multiplying thecomponent in the respective multiplier of multiplier bank 874 with therespective weighting factor w₁, w₂, . . . , w_(L) from the AVCcorresponding to the delayed signal component. All weighted signalcomponents are summed in the adder 876 to give the combined multipathcomponent signal y(t).

[0222] The combined multipath component signal y(t) does not include thecorrection due to phase and frequency offset of the carrier signal. Thecorrection for the phase and frequency offset of the carrier signal ismade to y(t) by multiplying y(t) with carrier phase and frequencycorrection (derotation phasor) in multiplier 878. The phase andfrequency correction is produced by the AVC as described previously.FIG. 8d shows the correction as being applied before the despreadingcircuits 880, but alternate embodiments of the invention can apply thecorrection after the despreading circuits.

[0223] Method to Reduce Re-Acquisition Time with Virtual Location

[0224] One consequence of determining the difference in code phasebetween the locally generated pilot code sequence and a receivedspreading code sequence is that an approximate value for the distancebetween the base station and a subscriber unit can be calculated. If theSU has a relatively fixed position with respect to the RCS of the basestation, the uncertainty of received spreading code phase is reduced forsubsequent attempts at re-acquisition by the SU or RCS. The timerequired for the base station to acquire the access signal of a SU thathas gone “off-hook” contributes to the delay between the SU goingoff-hook and the receipt of a dial tone from the PSTN. For systems thatrequire a short delay, such as 150 msec for dial tone after off-hook isdetected, a method which reduces the acquisition and bearer channelestablishment time is desirable. One embodiment of the present inventionuses such a method of reducing re-acquisition by use of virtuallocating. Additional details of this technique are described in U.S.Patent Application entitled “VIRTUAL LOCATING OF A FIXED SUBSCRIBER UNITTO REDUCE RE-ACQUISITION TIME” filed on even date herewith andincorporated herein by reference.

[0225] The RCS acquires the SU CDMA signal by searching only thosereceived code phases corresponding to the largest propagation delay ofthe particular system. In other words, the RCS assumes that all SUs areat a predetermined, fixed distance from the RCS. The first time the SUestablishes a channel with the RCS, the normal search pattern isperformed by the RCS to acquire the access channel. The normal methodstarts by searching the code phases corresponding to the longestpossible delay, and gradually adjusts the search to the code phases withthe shortest possible delay. However, after the initial acquisition, theSU can calculate the delay between the RCS and the SU by measuring thetime difference between sending a short access message to the RCS andreceiving an acknowledgment message, and using the received Global Pilotchannel as a timing reference. The SU can also receive the delay valueby having the RCS calculate the round trip delay difference from thecode phase difference between the Global Pilot code generated at the RCSand the received assigned pilot sequence from the SU, and then sendingthe SU the value on a predetermined control channel. Once the round tripdelay is known to the SU , the SU may adjust the code phase of thelocally generated assigned pilot and spreading code sequences by addingthe delay required to make the SU appear to the RCS to be at thepredetermined fixed distance from the RCS. Although the method isexplained for the largest delay, a delay corresponding to anypredetermined location in the system can be used.

[0226] A second advantage of the method of reducing re-acquisition byvirtual locating is that a conservation in SU power use can be achieved.Note that a SU that is “Powered down” or in a sleep mode needs to startthe bearer channel acquisition process with a low transmit power leveland ramp-up power until the RCS can receive its signal in order tominimize interference with other users. Since the subsequentre-acquisition time is shorter, and because the SU's location isrelatively fixed in relation to the RCS, the SU can ramp-up transmitpower more quickly because the SU will wait a shorter period of timebefore increasing transmit power. The SU waits a shorter period becauseit knows, within a small error range, when it should receive a responsefrom the RCS if the RCS has acquired the SU signal.

The Spread Spectrum Communication System

[0227] The Radio Carrier Station (RCS)

[0228] The Radio Carrier Station (RCS) of the present invention acts asa central interface between the SU and the remote processing controlnetwork element, such as a Radio Distribution Unit (RDU). The interfaceto the RDU of the present embodiment follows the G.704 standard and aninterface according to a modified version of DECT V5.1, but the presentinvention can support any interface that can exchange call control andtraffic channels. The RCS receives information channels from the RDUincluding call control data, and traffic channel data such as, but notlimited to, 32 kb/s ADPCM, 64 kb/s PCM, and ISDN, as well as systemconfiguration and maintenance data. The RCS also terminates the CDMAradio interface bearer channels with SUs, which channels include bothcontrol data, and traffic channel data. In response to the call controldata from either the RDU or a SU, the RCS allocates traffic channels tobearer channels on the RF communication link and establishes acommunication connection between the SU and the telephone networkthrough an RDU.

[0229] As shown in FIG. 9, the RCS receives call control and messageinformation data into the MUXs 905, 906 and 907 through interface lines901, 902 and 903. Although E1 format is shown, other similartelecommunication formats can be supported in the same manner asdescribed below. The MUXs shown in FIG. 9 may be implemented usingcircuits similar to that shown in FIG. 10. The MUX shown in FIG. 10includes system clock signal generator 1001 consisting of phase lockedoscillators (not shown) which generate clock signals for the Line PCMhighway 1002 (which is part of PCM Highway 910), and high speed bus(HSB) 970; and the MUX Controller 1010 which synchronizes the systemclock 1001 to interface line 1004. It is contemplated that the phaselock oscillators can provide timing signals for the RCS in the absenceof synchronization to a line. The MUX Line Interface 1011 separates thecall control data from the message information data. Referring to FIG.9, each MUX provides a connection to the Wireless Access Controller(WAC) 920 through the PCM highway 910. The MUX controller 1010 alsomonitors the presence of different tones present in the informationsignal by means of tone detector 1030.

[0230] Additionally, the MUX Controller 1010 provides the ISDN D channelnetwork signaling locally to the RDU. The MUX line interface 1011, suchas a FALC 54, includes an E1 interface 1012 which consists of a transmitconnection pair (not shown) and a receive connection pair (not shown) ofthe MUX connected to the RDU or Central Office (CO) ISDN Switch at thedata rate of 2.048 Mbps. The transmit and receive connection pairs areconnected to the E1 interface 1012 which translates differentialtri-level transmit/receive encoded pairs into levels for use by theFramer 1015. The line interface 1011 uses internal phase-locked-loops(not shown) to produce E1-derived 2.048 MHz, and 4.096 MHz clocks aswell as an 8 KHz frame-sync pulse. The line interface can operate inclock-master or clock-slave mode. While the exemplary embodiment isshown as using an E1 Interface, it is contemplated that other types oftelephone lines which convey multiple calls may be used, for example, T1lines or lines which interface to a Private Branch Exchange (PBX).

[0231] The line interface framer 1015 frames the data streams byrecognizing the framing patterns on channel-1 (time-slot 0) of theincoming line, and inserts and extracts service bits, generates/checksline service quality information.

[0232] As long as a valid E1 signal appears at the E1 Interface 1012,the FALC 54, recovers a 2.048 MHz PCM clock signal from the E1 line.This clock, via System Clock 1001, is used system wide as a PCM HighwayClock signal. If the E1 Line fails, the FALC 54 continues to deliver aPCM Clock derived from an oscillator signal o(t) connected to the syncinput (not shown) of the FALC 54. This PCM Clock serves the RCS systemuntil another MUX with an operational E1 line assumes responsibility forgenerating the system clock signals.

[0233] The framer 1015 generates a Received Frame Sync Pulse, which inturn can be used to trigger the PCM Interface 1016 to transfer data ontothe line PCM Highway 1002 and into the RCS System for use by otherelements. Since all E1 lines are frame synchronized, all Line PCMHighways are also frame synchronized. From this 8 kHz PCM Sync pulse,the system clock signal generator 1001 of the MUX uses a Phase LockedLoop (not shown) to synthesize the PN×2 clock [e.g., 15.96 MHz)(W₀(t)].The frequency of this clock signal is different for differenttransmission bandwidths, as described in Table 7.

[0234] The MUX includes a MUX Controller 1010, such as a 25 MHz QuadIntegrated Communications Controller, containing a microprocessor 1020,program memory 1021, and Time Division Multiplexer (TDM) 1022. The TDM1022 is coupled to receive the signal provided by the Framer 1015, andextracts information placed in time slots 0 and 16. The extractedinformation governs how the MUX controller 1010 processes the LinkAccess Protocol-D (LAPD) data link. The call control and bearermodification messages, such as those defined as V5.1 Network layermessages, are either passed to the WAC, or used locally by the MUXcontroller 1010.

[0235] The RCS Line PCM Highway 1002 is connected to and originates withthe Framer 1015 through PCM Interface 1016, and comprises of a 2.048 MHzstream of data in both the transmit and receive direction. The RCS alsocontains a High Speed Bus (HSB) 970 which is the communication linkbetween the MUX, WAC, and MIUs. The HSB 970 supports a data rate of, forexample, 100 Mbit/sec. Each of the MUX, WAC, and MIU access the HSBusing arbitration. The RCS of the present invention also can includeseveral MUXs requiring one board to be a “master” and the rest “slaves”.Details on the implementation of the HSB may be found in a U.S. patentapplication entitled PARALLEL PACKETIZED INTERMODULE ARBITRATED HIGHSPEED CONTROL AND DATA BUS, filed on even date herewith, which is herebyincorporated by reference.

[0236] Referring to FIG. 9, the Wireless Access Controller (WAC) 920 isthe RCS system controller which manages call control functions andinterconnection of data streams between the MUXs 905, 906, 907, ModemInterface Units (MIUs) 931, 932, 933. The WAC 920 also controls andmonitors other RCS elements such as the VDC 940, RF 950, and PowerAmplifiers 960. The WAC 920 as shown in FIG. 11, allocates bearerchannels to the modems on each MIU 931, 932, 933 and allocates themessage data on line PCM Highway 910 from the MUXs 905, 906, 907 to themodems on the MIUs 931, 932, 933. This allocation is made through theSystem PCM Highway 911 by means of a time slot interchange on the WAC920. If more than one WAC is present for redundancy purposes, the WACsdetermines the Master-Slave relationship with a second WAC. The WAC 920also generates messages and paging information responsive to callcontrol signals from the MUXs 905, 906, 907 received from a remoteprocessor, such as an RDU; generates Broadcast Data which is transmittedto the MIU master modem 934; and controls the generation by the MIU MM934 of the Global system Pilot spreading code sequence. The WAC 920 alsois connected to an external Network Manager (NM) 980 for craftperson oruser access.

[0237] Referring to FIG. 11, the WAC includes a time-slot interchanger(TSI) 1101 which transfers information from one time slot in a Line PCMHighway or System PCM Highway to another time slot in either the same ordifferent Line PCM Highway or System PCM Highway. The TSI 1101 isconnected to the WAC controller 1111 of FIG. 11 which controls theassignment or transfer of information from one time slot to another timeslot and stores this information in memory 1120. The exemplaryembodiment of the invention has four PCM Highways 1102, 1103, 1104, 1105connected to the TSI. The WAC also is connected to the HSB 970, throughwhich WAC communicates to a second WAC (not shown), to the MUXs and tothe MIUs.

[0238] Referring to FIG. 11, the WAC 920 includes a WAC controller 1111employing, for example, a microprocessor 1112, such as a MotorolaMC68040 and a communications processor 1113, such as the MotorolaMC68360 QUICC communications processor, and a clock oscillator 1114which receives a clock synch signal wo(t) from the system clockgenerator. The clock generator is located on a MUX (not shown) toprovide timing to the WAC controller 1111. The WAC controller 1111 alsoincludes memory 1120 including Flash Prom 1121 and SRAM memory 1122. TheFlash Prom 1121 contains the program code for the WAC controller 1111,and is reprogrammable for new software programs downloaded from anexternal source. The SRAM 1122 is provided to contain the temporary datawritten to and read from memory 1120 by the WAC controller 1111.

[0239] A low speed bus 912 is connected to the WAC 920 for transferringcontrol and status signals between the RF Transmitter/Receiver 950, VDC940, RF 950 and Power Amplifier 960 as shown in FIG. 9. The controlsignals are sent from the WAC 920 to enable or disable the RFTransmitters/Receiver 950 or Power amplifier 960, and the status signalsare sent from the RF Transmitters/Receiver 950 or Power amplifier 960 tomonitor the presence of a fault condition.

[0240] Referring to FIG. 9, the exemplary RCS contains at least one MIU931, which is shown in FIG. 12 and now described in detail. The MIU ofthe exemplary embodiment includes six CDMA modems, but the invention isnot limited to this number of modems. The MIU includes a System PCMHighway 1201 connected to each of the CDMA Modems 1210, 1211, 1212, 1215through a PCM Interface 1220, a Control Channel Bus 1221 connected toMIU controller 1230 and each of the CDMA Modems 1210, 1211, 1212, 1213,an MIU clock signal generator (CLK) 1231, and a modem output combiner1232. The MIU provides the RCS with the following functions: the MIUcontroller receives CDMA Channel Assignment Instructions from the WACand assigns a modem to a user information signal which is applied to theline interface of the MUX and a modem to receive the CDMA channel fromthe SU; it also combines the CDMA Transmit Modem Data for each of theMIU CDMA modems; multiplexes I and Q transmit message data from the CDMAmodems for transmission to the VDC; receives Analog I and Q receivemessage data from the VDC; distributes the I and Q data to the CDMAmodems; transmits and receives digital AGC Data; distributes the AGCdata to the CDMA modems; and sends MIU Board Status and MaintenanceInformation to the WAC 920.

[0241] The MIU controller 1230 of the exemplary embodiment of thepresent invention contains one communication microprocessor 1240, suchas the MC68360 “QUICC” Processor, and includes a memory 1242 having aFlash Prom memory 1243 and a SRAM memory 1244. Flash Prom 1243 isprovided to contain the program code for the Microprocessors 1240, andthe memory 1243 is downloadable and reprogrammable to support newprogram versions. SRAM 1244 is provided to contain the temporary dataspace needed by the MC68360 Microprocessor 1240 when the MIU controller1230 reads or writes data to memory

[0242] The MIU CLK circuit 1231 provides a timing signal to the MIUcontroller 1230, and also provides a timing signal to the CDMA modems.The MIU CLK circuit 1231 receives and is synchronized to the systemclock signal wo(t). The controller clock signal generator 1213 alsoreceives and synchronizes to the spreading code clock signal pn(t) whichis distributed to the CDMA modems 1210, 1211, 1212, 1215 from the MUX.

[0243] The RCS of the present embodiment includes a System Modem 1210contained on one MIU. The System Modem 1210 includes a Broadcastspreader (not shown) and a Pilot Generator (not shown). The BroadcastModem provides the broadcast information used by the exemplary system,and the broadcast message data is transferred from the MIU controller1230 to the System Modem 1210. The System Modem also includes fouradditional modems (not shown) which are used to transmit the signals CT1through CT4 and AX1 through AX4. The System Modem 1210 providesunweighted I and Q Broadcast message data signals which are applied tothe VDC. The VDC adds the Broadcast message data signal to the MIU CDMAModem Transmit Data of all CDMA modems 1210, 1211, 1212, 1215, and theGlobal Pilot signal.

[0244] The Pilot Generator (PG) 1250 provides the Global Pilot signalwhich is used by the present invention, and the Global Pilot signal isprovided to the CDMA modems 1210, 1211, 1212, 1215 by the MIU controller1230. However, other embodiments of the present invention do not requirethe MIU controller to generate the Global Pilot signal, but include aGlobal Pilot signal generated by any form of CDMA Code Sequencegenerator. In the described embodiment of the invention, the unweightedI and Q Global Pilot signal is also sent to the VDC where it is assigneda weight, and added to the MIU CDMA Modem transmit data and Broadcastmessage data signal.

[0245] System timing in the RCS is derived from the E1 interface. Thereare four MUXs in an RCS, three of which (905, 906 and 907) are shown inFIG. 9. Two MUXs are located on each chassis. One of the two MUXs oneach chassis is designated as the master, and one of the masters isdesignated as the system master. The MUX which is the system masterderives a 2.048 Mhz PCM clock signal from the E1 interface using a phaselocked loop (not shown). In turn, the system master MUX divides the2.048 Mhz PCM clock signal in frequency by 16 to derive a 128 KHzreference clock signal. The 128 KHz reference clock signal isdistributed from the MUX that is the system master to all the otherMUXs. In turn, each MUX multiplies the 128 KHz reference clock signal infrequency to synthesize the system clock signal which has a frequencythat is twice the frequency of the PN-clock signal. The MUX also dividesthe 128 KHz clock signal in frequency by 16 to generate the 8 KHz framesynch signal which is distributed to the MIUs. The system clock signalfor the exemplary embodiment has a frequency of 11.648 Mhz for a 7 MHzbandwidth CDMA channel Each MUX also divides the system clock signal infrequency by 2 to obtain the PN-clock signal and further divides thePN-clock signal in frequency by 29 877 120 (the PN sequence length) togenerate the PN-synch signal which indicates the epoch boundaries. ThePN-synch signal from the system master MUX is also distributed to allMUXs to maintain phase alignment of the internally generated clocksignals for each MUX. The PN-synch signal and the frame synch signal arealigned. The two MUXs that are designated as the master MUXs for eachchasis then distribute both the system clock signal and the PN-clocksignal to the MIUs and the VDC.

[0246] The PCM Highway Interface 1220 connects the System PCM Highway911 to each CDMA Modem 1210, 1211, 1212, 1215. The WAC controllertransmits Modem Control information, including traffic message controlsignals for each respective user information signal, to the MIUcontroller 1230 through the HSB 970. Each CDMA Modem 1210, 1211, 1212,1215 receives a traffic message control signal, which includes signalinginformation, from the MIU controller 1111. Traffic message controlsignals also include call control (CC) information and spreading codeand despreading code sequence information.

[0247] The MIU also includes the Transmit Data Combiner 1232 which addsweighted CDMA modem transmit data including In-phase (I) and Quadrature(Q) modem transmit data from the CDMA modems 1210, 1211, 1212, 1215 onthe MIU. The I modem transmit data is added separately from the Q modemtransmit data. The combined I and Q modem transmit data output signal ofthe Transmit Data Combiner 1232 is applied to the I and Q multiplexer1233 that creates a single CDMA transmit message channel composed of theI and Q modem transmit data multiplexed into a digital data stream.

[0248] The Receiver Data Input Circuit (RDI) 1234 receives the AnalogDifferential I and Q Data from the Video Distribution Circuit (VDC) 940shown in FIG. 9 and distributes Analog Differential I and Q Data to eachof the CDMA Modems 1210, 1211, 1212, 1215 of the MIU. The Automatic GainControl Distribution Circuit (AGC) 1235 receives the AGC Data signalfrom the VDC and distributes the AGC Data to each of the CDMA Modems ofthe MIU. The TRL circuit 1233 receives the Traffic lights informationand similarly distributes the Traffic light data to each of the Modems1210, 1211, 1212, 1215.

[0249] The CDMA Modem

[0250] The CDMA modem provides for generation of CDMA spreading codesequences and synchronization between transmitter and receiver. It alsoprovides four full duplex channels (TR0, TR1, TR2, TR3) programmable to64, 32, 16, and 8 ksym/sec. each, for spreading and transmission at aspecific power level. The CDMA modem measures the received signalstrength to allow Automatic Power Control, it generates and transmitspilot signals, and encodes and decodes using the signal for forwarderror correction (FEC). The modem in an SU also performs transmitterspreading code pulse shaping using an FIR filter. The CDMA modem is alsoused by the Subscriber Unit (SU), and in the following discussion thosefeatures which are used only by the SU are distinctly pointed out. Theoperating frequencies of the CDMA modem are given in Table 10. TABLE 10Operating Frequencies Bandwidth Chip Rate Symbol Rate Gain (MHz) (MHz)(KHz) (Chips/Symbol) 7 5.824 64  91 10 8.320 64 130 10.5 8.512 64 133 1411.648 64 182 15 12.480 64 195

[0251] Each CDMA modem 1210, 1211, 1212, 1215 of FIG. 12, and as shownin FIG. 13, is composed of a transmit section 1301 and a receive section1302. Also included in the CDMA modem is a control center 1303 whichreceives control messages CNTRL from the external system. These messagesare used, for example, to assign particular spreading codes, activatethe spreading or despreading, or to assign transmission rates. Inaddition, the CDMA modem has a code generator means 1304 used togenerate the various spreading and despreading codes used by the CDMAmodem. The transmit section 1301 is for transmitting the inputinformation and control signals m_(i)(t), i=1, 2, . . . I asspread-spectrum processed user information signals sc_(j)(t), j=1, 2, .. . J. The transmit section 1301 receives the global pilot code from thecode generator 1304 which is controlled by the control means 1303. Thespread spectrum processed user information signals are ultimately addedto other similar processed signals and transmitted as CDMA channels overthe CDMA RF forward message link, for example to the SUs. The receivesection 1302 receives CDMA channels as r(t) and despreads and recoversthe user information and control signals rc_(k)(t), k=1, 2, . . . Ktransmitted over the CDMA RF reverse message link, for example to theRCS from the SUs.

[0252] CDMA Modem Transmitter Section

[0253] Referring to FIG. 14, the code generator means 1304 includesTransmit Timing Control Logic 1401 and spreading code PN-Generator 1402,and the Transmit Section 1301 includes Modem Input Signal Receiver(MISR) 1410, Convolution Encoders 1411, 1412, 1413, 1414, Spreaders1420, 1421, 1422, 1423, 1424, and Combiner 1430. The Transmit Section1301 receives the message data channels MESSAGE, convolutionally encodeseach message data channel in the respective convolutional encoder 1411,1412, 1413, 1414, modulates the data with random spreading code sequencein the respective spreader 1420, 1421, 1422, 1423, 1424, and combinesmodulated data from all channels, including the pilot code received inthe described embodiment from the code generator, in the combiner 1430to generate I and Q components for RF transmission. The TransmitterSection 1301 of the present embodiment supports four (TR0, TR1, TR2,TR3) 64, 32, 16, 8 kb/s programmable channels. The message channel datais a time multiplexed signal received from the PCM highway 1201 throughPCM interface 1220 and input to the MISR 1410.

[0254]FIG. 15 is a block diagram of an exemplary MISR 1410. For theexemplary embodiment of the present invention, a counter is set by the 8KHz frame synchronization signal MPCMSYNC and is incremented by 2.048MHz MPCMCLK from the timing circuit 1401. The counter output is comparedby comparator 1502 against TRCFG values corresponding to slot timelocation for TR0, TR1, TR2, TR3 message channel data; and the TRCFGvalues are received from the MIU Controller 1230 in MCTRL. Thecomparator sends count signal to the registers 1505, 1506, 1507 and 1508which clocks message channel data into buffers 1510, 1511, 1512, 1513using the TXPCNCLK timing signal derived from the system clock. Themessage data is provided from the signal MSGDAT from the PCM highwaysignal MESSAGE when enable signals TR0EN, TR1EN, TR2EN and TR3EN fromTiming Control Logic 1401 are active. In further embodiments, MESSAGEmay also include signals that enable registers depending upon anencryption rate or data rate. If the counter output is equal to one ofthe channel location addresses, the specified transmit message data inregisters 1510, 1511, 1512, 1513 are input to the convolutional encoders1411, 1412, 1413, 1414 shown in FIG. 14.

[0255] The convolutional encoder enables the use of Forward ErrorCorrection (FEC) techniques, which are well known in the art. FECtechniques depend on introducing redundancy in generation of data inencoded form. Encoded data is transmitted and the redundancy in the dataenables the receiver decoder device to detect and correct errors. Oneembodiment of the present invention employs convolutional encoding.Additional data bits are added to the data in the encoding process andare the coding overhead. The coding rate is expressed as the ratio ofdata bits transmitted to the total bits (code data+redundant data)transmitted and is called the rate “R” of the code.

[0256] Convolution codes are codes where each code bit is generated bythe convolution of each new uncoded bit with a number of previouslycoded bits. The total number of bits used in the encoding process isreferred to as the constraint length, “K”, of the code. In convolutionalcoding, data is clocked into a shift register of K bits length so thatan incoming bit is clocked into the register, and it and the existingK−1 bits are convolutionally encoded to create a new symbol. Theconvolution process consists of creating a symbol consisting of amodule-2 sum of a certain pattern of available bits, always includingthe first bit and the last bit in at least one of the symbols.

[0257]FIG. 16 shows the block diagram of a K=7, R=½ convolution encodersuitable for use as the encoder 1411 shown in FIG. 14. This circuitencodes the TR0 Channel as used in one embodiment of the presentinvention. Seven-Bit Register 1601 with stages Q1 through Q7 uses thesignal TXPNCLK to clock in TR0 data when the TR0EN signal is asserted.The output value of stages Q1, Q2, Q3, Q4, Q6, and Q7 are each combinedusing EXCLUSIVE-OR Logic 1602, 1603 to produce respective I and Qchannel FEC data for the TR0 channel FECTR0DI and FECTR0DQ.

[0258] Two output symbol streams FECTR0DI and FECTR0DQ are generated.The FECTR0DI symbol stream is generated by EXCLUSIVE OR Logic 1602 ofshift register outputs corresponding to bits 6, 5, 4, 3, and 0, (Octal171) and is designed as In phase component “I” of the transmit messagechannel data. The symbol stream FECTR0DQ is likewise generated byEXCLUSIVE-OR logic 1603 of shift register outputs from bits 6, 4 3, 1and 0, (Octal 133) and is designated as Quadrature component “Q” of thetransmit message channel data. Two symbols are transmitted to representa single encoded bit creating the redundancy necessary to enable errorcorrection to take place on the receiving end.

[0259] Referring to FIG. 14, the shift enable clock signal for thetransmit message channel data is generated by the Control Timing Logic1401. The convolutionally encoded transmit message channel output datafor each channel is applied to the respective spreader 1420, 1421, 1422,1423, 1424 which multiplies the transmit message channel data by itspreassigned spreading code sequence from code generator 1402. Thisspreading code sequence is generated by control 1303 as previouslydescribed, and is called a random pseudonoise signature sequence(PN-code).

[0260] The output signal of each spreader 1420, 1421, 1422, 1423, 1424is a spread transmit data channel. The operation of the spreader is asfollows: the spreading of channel output (I+jQ) multiplied by a randomsequence (PNI+jPNQ) yields the In-phase component I of the result beingcomposed of (I xor PNI) and (-Q xor PNQ). Quadrature component Q of theresult is (Q xor PNI) and (I xor PNQ). Since there is no channel datainput to the pilot channel logic (I=1, Q values are prohibited), thespread output signal for pilot channels yields the respective sequencesPNI for I component and PNQ for Q component.

[0261] The combiner 1430 receives the I and Q spread transmit datachannels and combines the channels into an I modem transmit data signal(TXIDAT) and a Q modem transmit data signal (TXQDAT). The I-spreadtransmit data and the Q spread transmit data are added separately.

[0262] For an SU, the CDMA modem Transmit Section 1301 includes the FIRfilters to receive the I and Q channels from the combiner to providepulse shaping, close-in spectral control and x/sin(x) correction for thetransmitted signal. Separate but identical FIR filters receive the I andQ spread transmit data streams at the chipping rate, and the outputsignal of each of the filters is at twice the chipping rate. Theexemplary FIR filters are 28 tap even symmetrical filters, whichupsample (interpolate) by 2. The upsampling occurs before the filtering,so that 28 taps refers to 28 taps at twice the chipping rate, and theupsampling is accomplished by setting every other sample a zero.Exemplary coefficients are shown in Table 11. TABLE 11 CoefficientValues Coeff. No.: 0 1 2 3 4 5 6 7 8 9 10 11 12 13 Value: 3 −11 −34 −2219 17 −32 −19 52 24 −94 −31 277 468 Coeff. No.: 14 15 16 17 18 19 20 2122 24 25 26 27 Value: 277 −31 −94 24 52 −19 −32 17 19 −22 −34 −11 3

[0263] CDMA Modem Receiver Section

[0264] Referring to FIGS. 9 and 12, the RF receiver 950 of the presentembodiment accepts analog input I and Q CDMA channels, which aretransmitted to the CDMA modems 1210, 1211, 1212, 1215 through the MIUs931, 932, 933 from the VDC 940. These I and Q CMDA channel signals aresampled by the CDMA modem receive section 1302 (shown in FIG. 13) andconverted to I and Q digital receive message signal using an Analog toDigital (A/D) converter 1730, shown in FIG. 17. The sampling rate of theA/D converter of the exemplary embodiment of the present invention isequivalent to the despreading code rate. The I and Q digital receivemessage signals are then despread with correlators using six differentcomplex spreading code sequences corresponding to the despreading codesequences of the four channels (TR0, TR1, TR2, TR3), APC information andthe pilot code.

[0265] Time synchronization of the receiver to the received signal isseparated into two phases; there is an initial acquisition phase andthen a tracking phase after the signal timing has been acquired. Theinitial acquisition is done by shifting the phase of the locallygenerated pilot code sequence relative to the received signal andcomparing the output of the pilot despreader to a threshold. The methodused is called sequential search. Two thresholds (match and dismiss) arecalculated from the auxiliary despreader. Once the signal is acquired,the search process is stopped and the tracking process begins. Thetracking process maintains the code generator 1304 (shown in FIGS. 13and 17) used by the receiver in synchronization with the incomingsignal. The tracking loop used is the Delay-Locked Loop (DLL) and isimplemented in the acquisition & track 1701 and the IPM 1702 blocks ofFIG. 17.

[0266] In FIG. 13, the modem controller 1303 implements the Phase LockLoop (PLL) as a software algorithm in SW PLL logic 1724 of FIG. 17 thatcalculates the phase and frequency shift in the received signal relativeto the transmitted signal. The calculated phase shifts are used toderotate the phase shifts in rotate and combine blocks 1718, 1719, 1720,1721 of the multipath data signals for combining to produce outputsignals corresponding to receive channels TR0′, TR1′, TR2′, TR3′. Thedata is then Viterbi decoded in Viterbi Decoders 1713, 1714, 1715, 1716to remove the convolutional encoding in each of the received messagechannels.

[0267]FIG. 17 indicates that the Code Generator 1304 provides the codesequences Pn_(i)(t), i=1, 2, . . . I used by the receive channeldespreaders 1703, 1704, 1705, 1706, 1707, 1708, 1709. The code sequencesgenerated are timed in response to the SYNK signal of the system clocksignal and are determined by the CCNTRL signal from the modem controller1303 shown in FIG. 13. Referring to FIG. 17, the CDMA modem receiversection 1302 includes Adaptive Matched Filter (AMF) 1710, Channeldespreaders 1703, 1704, 1705, 1706, 1707, 1708, 1709, Pilot AVC 1711,Auxiliary AVC 1712, Viterbi decoders 1713, 1714, 1715, 1716, Modemoutput interface (MOI) 1717, Rotate and Combine logic 1718, 1719, 1720,1721, AMF Weight Generator 1722, and Quantile Estimation logic 1723.

[0268] In another embodiment of the invention, the CDMA modem receiveralso includes a Bit error Integrator to measure the BER of the channeland idle code insertion logic between the Viterbi decoders 1713, 1714,1715, 1716 and the MOI 1717 to insert idle codes in the event of loss ofthe message data.

[0269] The Adaptive Matched Filter (AMF) 1710 resolves multipathinterference introduced by the air channel. The exemplary AMF 1710 usesan 11 stage complex FIR filter as shown in FIG. 18. The received I and Qdigital message signals are received at the register 1820 from the A/D1730 of FIG. 17 and are multiplied in multipliers 1801, 1802, 1803,1810, 1811 by I and Q channel weights W1 to W11 received from AMF weightgenerator 1722 of FIG. 17. In the exemplary embodiment, the A/D 1730provides the I and Q digital receive message signal data as 2'scomplement values, 6 bits for I and 6 bits for Q which are clockedthrough an 11 stage shift register 1820 responsive to the receivespreading-code clock signal RXPNCLK. The signal RXPNCLK is generated bythe timing section 1401 of code generation logic 1304. Each stage of theshift register is tapped and complex multiplied in the multipliers 1801,1802, 1803, 1810, 1811 by individual (6-bit I and 6-bit Q) weight valuesto provide 11 tap-weighted products which are summed in adder 1830, andlimited to 7-bit I and 7-bit Q values.

[0270] The CDMA modem receive section 1302 (shown in FIG. 13) providesindependent channel despreaders 1703, 1704, 1705, 1706, 1707, 1708, 1709(shown in FIG. 17) for despreading the message channels. The describedembodiment despreads 7 message channels, each despreader accepting a1-bit I b 1-bit Q despreading code signal to perform a complexcorrelation of this code against a 8-bit I by 8-bit Q data input. The 7despreaders correspond to the 7 channels: Traffic Channel 0 (TR0′),TR1′, TR2′, TR3′, AUX (a spare channel), Automatic Power Control (APC)and pilot (PLT).

[0271] The Pilot AVC 1711 shown in FIG. 19 receives the I and Q PilotSpreading code sequence values PCI and PCQ into shift register 1920responsive to the timing signal RXPNCLK, and includes 11 individualdespreaders 1901 through 1911 each correlating the I and Q digitalreceive message signal data with a one chip delayed version of the samepilot code sequence. Signals OE1, OE2, . . . OE11 are used by the modemcontrol 1303 to enable the despreading operation. The output signals ofthe despreaders are combined in combiner 1920 forming correlation signalDSPRDAT of the Pilot AVC 1711, which is received by the ACQ & Tracklogic 1701 (shown in FIG. 17), and ultimately by modem controller 1303(shown in FIG. 13). The ACQ & Track logic 1701 uses the correlationsignal value to determine if the local receiver is synchronized with itsremote transmitter.

[0272] The Auxiliary AVC 1712 also receives the I and Q digital receivemessage signal data and, in the described embodiment, includes fourseparate despreaders 2001, 2002, 2003, 2004 as shown in FIG. 20. Eachdespreader receives and correlates the I and Q digital receive messagedata with delayed versions of the same despreading code sequence PARIand PARQ which are provided by code generator 1304 input to andcontained in shift register 2020. The output signals of the despreaders2001, 2002, 2003, 2004 are combined in combiner 2030 which providesnoise correlation signal ARDSPRDAT. The auxiliary AVC spreading codesequence does not correspond to any transmit spreading code sequence ofthe system. Signals OE1, OE2, . . . OE4 are used by the modem control1303 to enable the despreading operation. The Auxiliary AVC 1712provides a noise correlation signal ARDSPRDAT from which quantileestimates are calculated by the Quantile estimator 1733, and provides anoise level measurement to the ACQ & Track logic 1701 (shown in FIG. 17)and modem controller 1303 (shown in FIG. 13).

[0273] Each despread channel output signal corresponding to the receivedmessage channels TR0′, TR1′, TR2′, and TR3′is input to a correspondingViterbi decoder 1713, 1714, 1715, 1716 shown in FIG. 17 which performsforward error correction on convolutionally encoded data. The Viterbidecoders of the exemplary embodiment have a constraint length of K=7 anda rate of R=½. The decoded despread message channel signals aretransferred from the CDMA modem to the PCM Highway 1201 through the MOI1717. The operation of the MOI is essentially the same as the operationof the MISR of the transmit section 1301 (shown in FIG. 13) except inreverse.

[0274] The CDMA modem receiver section 1302 implements several differentalgorithms during different phases of the acquisition, tracking anddespreading of the receive CDMA message signal.

[0275] When the received signal is momentarily lost (or severelydegraded) the idle code insertion algorithm inserts idle codes in placeof the lost or degraded receive message data to prevent the user fromhearing loud noise bursts on a voice call. The idle codes are sent tothe MOI 1717 (shown in FIG. 17) in place of the decoded message channeloutput signal from the Viterbi decoders 1713, 1714, 1715, 1716. The idlecode used for each traffic channel is programmed by the Modem Controller1303 by writing the appropriate pattern IDLE to the MOI, which in thepresent embodiment is a 8 bit word for a 64 kb/s stream, 4 bit word fora 32 kb/s stream.

[0276] Modem Algorithms for Acquisition and Tracking of Received PilotSignal

[0277] The acquisition and tracking algorithms are used by the receiverto determine the approximate code phase of a received signal,synchronize the local modem receiver despreaders to the incoming pilotsignal, and track the phase of the locally generated pilot code sequencewith the received pilot code sequence. Referring to FIGS. 13 and 17, thealgorithms are performed by the Modem controller 1303, which providesclock adjust signals to code generator 1304. These adjust signals causethe code generator for the despreaders to adjust locally generated codesequences in response to measured output values of the Pilot Rake 1711and Quantile values from quantile estimators 1723B. Quantile values arenoise statistics measured from the In-phase and Quadrature channels fromthe output values of the AUX Vector Correlator 1712 (shown in FIG. 17).Synchronization of the receiver to the received signal is separated intotwo phases; an initial acquisition phase and a tracking phase. Theinitial acquisition phase is accomplished by clocking the locallygenerated pilot spreading code sequence at a higher or lower rate thanthe received signal's spreading code rate, sliding the locally generatedpilot spreading code sequence and performing sequential probabilityratio test (SPRT) on the output of the Pilot Vector correlator 1711. Thetracking phase maintains the locally generated spreading code pilotsequence in synchronization with the incoming pilot signal. Details ofthe quantile estimators 1723B may be found in U.S. patent applicationSer. No. 08/218,198 entitled “ADAPTIVE POWER CONTR0L FOR A SPREADSPECTRUM COMMUNICATIONS SYSTEM” which is incorporated by referenceherein for its teachings on adaptive power control systems.

[0278] The SU cold acquisition algorithm is used by the SU CDMA modemwhen it is first powered up, and therefore has no knowledge of thecorrect pilot spreading code phase, or when an SU attempts to reacquiresynchronization with the incoming pilot signal but has taken anexcessive amount of time. The cold acquisition algorithm is divided intotwo sub-phases. The first subphase consists of a search over the length233415 code used by the FBCH. Once this sub-code phase is acquired, thepilot's 233415×128 length code is known to within an ambiguity of 128possible phases. The second subphase is a search of these remaining 128possible phases. In order not to lose synch with the FBCH, in the secondphase of the search, it is desirable to switch back and forth betweentracking of the FBCH code and attempting acquisition of the pilot code.

[0279] The RCS acquisition of short access pilot (SAXPT) algorithm isused by an RCS CDMA modem to acquire the SAXPT pilot signal of an SU.Additional details of this technique are described in U.S. PatentApplication entitled “A METHOD OF CONTR0LLING INITIAL POWER RAMP-UP INCDMA SYSTEMS BY USING SHORT CODES” filed on even date herewith andherein incorporated by reference. The algorithm is a fast searchalgorithm because the SAXPT is a short code sequence of length N, whereN=chips/symbol, and ranges from 45 to 195, depending on the system'sbandwidth. The search cycles through all possible phases untilacquisition is complete.

[0280] The RCS acquisition of the long access pilot (LAXPT) algorithmbegins immediately after acquisition of SAXPT. The SU's code phase isknown within a multiple of a symbol duration, so in the exemplaryembodiment of the invention there may be 7 to 66 phases to search withinthe round trip delay from the RCS. This bound is a result of the SUpilot signal being synchronized to the RCS Global pilot signal.

[0281] The re-acquisition algorithm begins when loss of code lock (LOL)occurs. A Z-search algorithm is used to speed the process on theassumption that the code phase has not drifted far from where it was thelast time the system was locked. The RCS uses a maximum width of theZ-search windows bounded by the maximum round trip propagation delay.

[0282] The Pre-Track period immediately follows the acquisition orre-acquisition algorithms and immediately precedes the trackingalgorithm. Pre-track is a fixed duration period during which the receivedata provided by the modem is not considered valid. The Pre-Track periodallows other modem algorithms, such as those used by the ISW PLL 1724,ACQ & Tracking, AMF Weight GEN 1722, to prepare and adapt to the currentchannel. The Pre-Track period is two parts. The first part is the delaywhile the code tracking loop pulls in. The second part is the delaywhile the AMF tap weight calculations are performed by the AMF WeightGen 1722 to produce settled weighting coefficients. Also in the secondpart of the Pre-Track period, the carrier tracking loop is allowed topull in by the SW PLL 1724, and the scalar quantile estimates areperformed in the Quantile estimator 1723A.

[0283] The Tracking Process is entered after the Pre-Track period ends.This process is actually a repetitive cycle and is the only processphase during which receive data provided by the modem may be consideredvalid. The following operations are performed during this phase: AMF TapWeight Update, Carrier Tracking, Code Tracking, Vector Quantile Update,Scalar Quantile Update, Code Lock Check, Derotation and Symbol Summing,and Power Control (forward and reverse)

[0284] If LOL is detected, the modem receiver terminates the Trackalgorithm and automatically enters the reaquisition algorithm. In theSU, a LOL causes the transmitter to be shut down. In the RCS, LOL causesforward power control to be disabled with the transmit power heldconstant at the level immediately prior to loss of lock. It also causesthe return power control information being transmitted to assume a010101 . . . pattern, causing the SU to hold its transmit powerconstant. This can be performed using the signal lock check functionwhich generates the reset signal to the acquisition and tracking circuit1701.

[0285] Two sets of quantile statistics are maintained, one by Quantileestimator 1723B and the other by the scalar Quantile Estimator 1723A.Both are used by the modem controller 1303. The first set is the“vector” quantile information, so named because it is calculated fromthe vector of four complex values generated by the AUX AVC receiver1712. The second set is the scalar quantile information, which iscalculated from the single complex value AUX signal that is output fromthe AUX Despreader 1707. The two sets of information represent differentsets of noise statistics used to maintain a pre-determined Probabilityof False Alarm (P_(fa)). The vector quantile data is used by theacquisition and reaquisition algorithms implemented by the modemcontroller 1303 to determine the presence of a received signal in noise,and the scalar quantile information is used by the code lock checkalgorithm.

[0286] For both the vector and scalar cases, quantile informationconsists of calculated values of lambda0 through lambda2, which areboundary values used to estimate the probability distribution function(p.d.f) of the despread receive signal and determine whether the modemis locked to the PN code. The Aux_Power value used in the followingC-subroutine is the magnitude squared of the AUX signal output of thescalar correlator array for the scalar quantiles, and the sum of themagnitudes squared for the vector case. In both cases the quantiles arethen calculated using the following C-subroutine: for (n = 0; n < 3;n++) {   lambda [n] += (lambda [n] < Aux_Power) ? CG[n] : GM[n];   }

[0287] where CG[n] are positive constants and GM[n] are negativeconstants (different values are used for scalar and vector quantiles).

[0288] During the acquisition phase, the search of the incoming pilotsignal with the locally generated pilot code sequence employs a seriesof sequential tests to determine if the locally generated pilot code hasthe correct code phase relative to the received signal. The searchalgorithms use the Sequential Probability Ratio Test (SPRT) to determinewhether the received and locally generated code sequences are in phase.The speed of acquisition is increased by parallelism resulting fromhaving a multi-fingered receiver. For example, in the describedembodiment of the invention the main Pilot Rake 1711 has a total of 11fingers representing a total phase period of 11 chip periods. Foracquisition 8 separate sequential probability ratio tests (SPRTs) areimplemented, with each SPRT observing a 4 chip window. Each window isoffset from the previous window by one chip, and in a search sequenceany given code phase is covered by 4 windows. If all 8 of the SPRT testsare rejected, then the set of windows is moved by 8 chips. If any of theSPRT's is accepted, then the code phase of the locally generated pilotcode sequence is adjusted to attempt to center the accepted SPRT's phasewithin the Pilot AVC. It is likely that more than one SPRT reaches theacceptance threshold at the same time. A table lookup is used cover all256 possible combinations of accept/reject and the modem controller usesthe information to estimate the correct center code phase within thePilot Rake 1711. Each SPRT is implemented as follows (all operationsoccur at 64 k symbol rate): Denote the fingers' output level values asI_Finger[n] and Q_Finger[n], where n=0 . . . 10 (inclusive, 0 isearliest (most advanced) finger), then the power of each window is:${{Power}\quad {{Window}\quad\lbrack i\rbrack}} = {\sum\limits_{n}\left( {{{I\_ Finger}^{2}\lbrack n\rbrack} + {{Q\_ Finger}^{2}\lbrack n\rbrack}} \right)}$

[0289] To implement the SPRT's the modem controller then performs foreach of the windows the following calculations which are expressed as apseudo-code subroutine: /* find bin for Power */ tmp = SIGMA[0]; for (k= 0; k < 3; k++) {   if (Power > lambda [k]) tmp = SIGMA[k+1]; }test_statistic += tmp; /* update statistic */ if(test_statistic >ACCEPTANCE_THRESHOLD)you've got ACQ; else if (test_statistic <DISMISSAL_THRESHOLD) {   forget this code phase; } else keep trying -get more statistics;

[0290] where lambda[k] are as defined in the above section on quantileestimation, and SIGMA[k], ACCEPTANCE_THRESHOLD and DISMISSAL_THRESHOLDare predetermined constants. Note that SIGMA[k] is negative for valuesfor low values of k, and positive for right values of k, such that theacceptance and dismissal thresholds can be constants rather than afunction of how many symbols worth of data have been accumulated in thestatistic.

[0291] The modem controller determines which bin delimited by the valuesof lambda[k] the Power level falls into which allows the modemcontroller to develop an approximate statistic.

[0292] For the present algorithm, the control voltage is formed asε=y^(T)By, where y is a vector formed from the complex valued outputvalues of the Pilot Vector correlator 1711, and B is a matrix consistingof the constant values pre-determined to maximize the operatingcharacteristics while minimizing the noise as described previously withreference to the Quadratic Detector.

[0293] To understand the operation of the Quadratic Detector, it isuseful to consider the following. A spread spectrum (CDMA) signal, s(t)is passed through a multipath channel with an impulse response h_(c)(t).The baseband spread signal is described by equation (30).$\begin{matrix}{{s(t)} = {\sum\limits_{i}{C_{i}{p\left( {t - {iT}_{c}} \right)}}}} & (30)\end{matrix}$

[0294] where C_(i) is a complex spreading code symbol, p(t) is apredefined chip pulse and T_(c) is the chip time spacing, whereT_(c)=1/R_(c) and R_(c) is the chip rate.

[0295] The received baseband signal is represented by equation (31)$\begin{matrix}{{r(t)} = {{\sum\limits_{i}{C_{i}{q\left( {t - {iT}_{c} - \tau} \right)}}} + {n(t)}}} & (31)\end{matrix}$

[0296] where q(t)=p(t)*h_(c)(t), τ is an unknown delay and n(t) isadditive noise. The received signal is processed by a filter, h_(R)(t),so the waveform, x(t), to be processed is given by equation (32).$\begin{matrix}{{{x(t)} = {{\sum\limits_{i}{C_{i}{f\left( {t - {iT}_{c} - \tau} \right)}}} + {z(t)}}}{{{where}\quad {f(t)}} = {{{q(t)}*{h_{R}(t)}\quad {and}\quad {z(t)}} = {{n(t)}*{{h_{R}(t)}.}}}}} & (32)\end{matrix}$

[0297] In the exemplary receiver, samples of the received signal aretaken at the chip rate, that is to say, 1/T_(c). These samples,x(mT_(c)+τ′), are processed by an array of correlators that compute,during the r^(th) correlation period, the quantities given by equation(33) $\begin{matrix}{v_{k}^{(r)} = {\sum\limits_{m = {rL}}^{{rL} + L - 1}{{x\left( {{mT}_{c} + \tau^{\prime}} \right)}C_{m + k}^{*}}}} & (33)\end{matrix}$

[0298] These quantities are composed of a noise component w_(k) ^((r))and a deterministic component y_(k) ^((r)) given by equation (34).

y _(k) ^((r)) =E[v _(k) ^((r)) ]=Lf(kT _(c)+τ′−τ)  (34)

[0299] In the sequel, the time index r may be suppressed for ease ofwriting, although it is to be noted that the function f(t) changesslowly with time.

[0300] The samples are processed to adjust the sampling phase, τ′, in anoptimum fashion for further processing by the receiver, such as matchedfiltering. This adjustment is described below. To simplify therepresentation of the process, it is helpful to describe it in terms ofthe function f(t+τ), where the time-shift, τ, is to be adjusted. It isnoted that the function f(t+τ) is measured in the presence of noise.Thus, it may be problematical to adjust the phase τ′ based onmeasurements of the signal f(t+τ). To account for the noise, thefunction v(t): v(t)=f(t)+m(t) is introduced, where the term m(t)represents a noise process. The system processor may be derived based onconsiderations of the function v(t).

[0301] The process is non-coherent and therefore is based on theenvelope power function |v(t+τ)|². The functional e(τ′) given inequation (35) is helpful for describing the process.

e(τ′)=∫_(−∞) ⁰ |v(t+τ′−τ)|² dt−∫ ₀ ^(∞) |v(t+τ′−τ)|² dt  (35)

[0302] The shift parameter is adjusted for e(τ′)=0, which occurs whenthe energy on the interval (−∞, τ′−τ] equals that on the interval [τ′−τ,∞). The error characteristic is monotonic and therefore has a singlezero crossing point. This is the desirable quality of the functional. Adisadvantage of the functional is that it is ill-defined because theintegrals are unbounded when noise is present. Nevertheless, thefunctional e(τ′) may be cast in the form given by equation (36).

e(τ′)=∫^(−∞) ^(∞) w(t)|v(t+τ′−τ)|² dt  (36)

[0303] where the characteristic function w(t) is equal to sgn(t), thesignum function.

[0304] To optimize the characteristic function w(t), it is helpful todefine a figure of merit, F, as set forth in equation (37).$\begin{matrix}{F = \frac{\left\lbrack \overset{\_}{{e\left( {\tau_{0}^{\prime} + T_{A}} \right)} - {e\left( {\tau_{0}^{\prime} - T_{A}} \right)}} \right\rbrack^{2}}{{VAR}\left\{ {e\left( \tau_{0}^{\prime} \right)} \right\}}} & (37)\end{matrix}$

[0305] The numerator of F is the numerical slope of the mean errorcharacteristic on the interval [−TA, TA] surrounding the tracked value,τ₀′. The statistical mean is taken with respect to the noise as well asthe random channel, h_(c)(t). It is desirable to specify a statisticalcharacteristic of the channel in order to perform this statisticalaverage. For example, the channel may be modeled as a Wide SenseStationary Uncorrelated Scattering (WSSUS) channel with impulse responseh_(c)(t) and a white noise process U(t) that has an intensity functiong(t) as shown in equation (38).

h _(c)(t)={square root}{square root over (g(t)U(t))}  (38)

[0306] The variance of e(τ) is computed as the mean square value of thefuctuation

e′(τ)=e(τ)−<e(τ)>  (39)

[0307] where <e(τ)> is the average of e(τ) with respect to the noise.

[0308] Optimization of the figure of merit F with respect to thefunction w(t) may be carried out using well-known Variational methods ofoptimization.

[0309] Once the optimal w(t) is determined, the resulting processor maybe approximated accurately by a quadratic sample processor which isderived as follows.

[0310] By the sampling theorem, the signal v(t), bandlimited to abandwidth W may be expressed in terms of its samples as shown inequation (40).

v(t)=Σv(k/W)sin c[(Wt−k)π]  (40)

[0311] substituting this expansion into equation (z+6) results in aninfinite quadratic form in the samples v(k/W+τ′−τ). Making theassumption that the signal bandwidth equals the chip rate allows the useof a sampling scheme that is clocked by the chip clock signal to be usedto obtain the samples. These samples, v_(k) are represented by equation(41).

v _(k) =v(kT _(c)+τ′−τ)  (41)

[0312] This assumption leads to a simplification of the implementation.It is valid if the aliasing error is small.

[0313] In practice, the quadratic form that is derived is truncated. Anexample normalized B matrix is given below in Table 12. For thisexample, an exponential delay spread profile g(t)=exp(−t/τ) is assumedwith τ equal to one chip. An aperture parameter T_(A) equal to one andone-half chips has also been assumed. The underlying chip pulse has araised cosine spectrum with a 20% excess bandwidth. TABLE 12 Example Bmatrix 0 0 0 0 0 0 0 0 0 0 0 0 0 −0.1 0 0 0 0 0 0 0 0 0 −0.1 0.22 0.19−0.19 0 0 0 0 0 0 0 0 0.19 1 0.45 −0.2 0 0 0 0 0 0 0 −0.19 0.45 0.990.23 0 0 0 0 0 0 0 0 −0.2 0.23 0 −0.18 0.17 0 0 0 0 0 0 0 0 −0.18 −0.87−0.42 0.18 0 0 0 0 0 0 0 0.17 −0.42 −0.92 −0.16 0 0 0 0 0 0 0 0 0.18−0.16 −0.31 0 0 0 0 0 0 0 0 0 0 0 −0.13 0 0 0 0 0 0 0 0 0 0 0 0

[0314] Code tracking is implemented via a loop phase detector that isimplemented as follows. The vector y is defined as a column vector whichrepresents the 11 complex output level values of the Pilot AVC 1711, andB denotes an 11×11 symmetric real valued coefficient matrix withpre-determined values to optimize performance with the non-coherentPilot AVC output values y. The output signal ε of the phase detector isgiven by equation (42):

e=y^(T)By  (42)

[0315] The following calculations are then performed to implement aproportional plus integral loop filter and the VCO:

x[n]=x[n−1]+βε

z[n]=z[n−1]+x[n]+αε

[0316] for β and α which are constants chosen from modeling the systemto optimize system performance for the particular transmission channeland application, and where x[n] is the loop filter's integrator outputvalue and z[n] is the VCO output value. The code phase adjustments aremade by the modem controller the following C-subroutine: if (z > zmx) {  delay phase 1/16 chip;   z −= zmax; } else if (z < −zmax) {   advancephase 1/16 chip;   z += zmax; }

[0317] A different delay phase could be used in the above pseudo-codeconsistant with the present invention.

[0318] The AMF Tap-Weight Update Algorithm of the AMF Weight Gen 1722occurs periodically to de-rotate and scale the phase of each fingervalue of the Pilot Rake 1711 by performing a complex multiplication ofthe Pilot AVC finger value with the complex conjugate of the currentoutput value of the carrier tracking loop and applying the product to alow pass filter and form the complex conjugate of the filter values toproduce AMF tap-weight values, which are periodically written into theAMF filters of the CDMA modem.

[0319] The lock check algorithm, shown in FIG. 17, is implemented by themodem controller 1303 performing SPRT operations on the output signal ofthe scalar correlator array. The SPRT technique is the same as that forthe acquisition algorithms, except that the acceptance and rejectionthresholds are changed to increase the probability of detection of lock.

[0320] Carrier tracking is accomplished via a second order loop thatoperates on the pilot output values of the scalar correlated array. Thephase detector output is the hard limited version of the quadraturecomponent of the product of the (complex valued) pilot output signal ofthe scalar correlated array and the VCO output signal. The loop filteris a proportional plus integral design. The VCO is a pure summation,accumulated phase error φ, which is converted to the complex phasor cosφ+j sin φ using a look-up table in memory.

[0321] The previous description of acquisition and tracking algorithmfocuses on a non-coherent method because the acquisition and trackingalgorithm described requires non-coherent acquisition following bynon-coherent tracking because during acquisition a coherent reference isnot available until the AMF, Pilot AVC, Aux AVC, and DPLL are in anequilibrium state. However, it is known in the art that coherenttracking and combining is always optimal because in non-coherenttracking and combining the output phase information of each Pilot AVCfinger is lost. Consequently, another embodiment of the inventionemploys a two step acquisition and tracking system, in which thepreviously described non-coherent acquisition and tracking algorithm isimplemented first, and then the algorithm switches to a coherenttracking method. The coherent combining and tracking method is similarto that described previously, except that the error signal tracked is ofthe form:

ε=y^(T)Ay  (43)

[0322] where y is defined as a column vector which represents the 11complex output level values of the Pilot AVC 1711, and A denotes an11×11 symmetric real valued coefficient matrix with pre-determinedvalues to optimize performance with the coherent Pilot AVC outputs y. Anexemplary A matrix is shown below. $\begin{matrix}{A = \begin{matrix}1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 1 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & {- 1} & 0 & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 0 & {- 1} & 0 & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {- 1} & 0 & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {- 1} & 0 \\0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {- 1}\end{matrix}} & (44)\end{matrix}$

[0323] Referring to FIG. 9, the Video Distribution Controller Board(VDC) 940 of the RCS is connected to each MIU 931, 932, 933 and the RFTransmitters/Receivers 950. The VDC 940 is shown in FIG. 21. The DataCombiner Circuitry (DCC) 2150 includes a Data Demultiplexer 2101, DataSummer 2102, FIR Filters 2103, 2104, and a Driver 2111. The DCC 2150 1)receives the weighted CDMA modem I and Q data signal MDAT from each ofthe MIUs, 931, 932, 933, 2) sums the I and Q data with the digitalbearer channel data from each MIU 931, 932, 933, 3) and sums the resultwith the broadcast data message signal BCAST and the Global Pilotspreading code GPILOT provided by the master MIU modem 1210, 4) bandshapes the summed signals for transmission, and 5) produces analog datasignal for transmission to the RF Transmitter/Receiver.

[0324] FIR Filters 2103, 2104 are used to modify the MIU CDMA Transmit Iand Q Modem Data before transmission. The WAC transfers FIR FilterCoefficient data through the Serial Port link 912 through the VDCController 2120 and to the FIR filters 2103, 2104. Each FIR Filter 2103,2104 is Configured separately. The FIR Filters 2103, 2104 employUp-Sampling to operate at twice the chip rate so zero data values aresent after every MIU CDMA Transmit Modem DATI and DATQ value to produceFTXI and FTXQ

[0325] The VDC 940 distributes the AGC signal AGCDATA from the AGC 1750of the MIUs 931, 932, 933 to the RF Transmitter/Receiver 950 through theDistribution interface (DI) 2110. The VDC DI 2110 receives data RXI andRXQ from the RF Transmitter/Receiver and distributes the signal asVDATAI and VDATAQ to MIUs 931, 932, 933.

[0326] Referring to FIG. 21, the VDC 940 also includes a VDC controller2120 which monitors status and fault information signals MIUSTAT fromMIUs and connects to the serial link 912 and HSBS 970 to communicatewith WAC 920 shown in FIG. 9. The VDC controller 2120 includes amicroprocessor, such as an Intel 8032 Microcontroller, an oscillator(not shown) providing timing signals, and memory (not shown). The VDCcontroller memory includes a Flash Prom (not shown) to contain thecontroller program code for the 8032 Microprocessor, and an SRAM (notshown) to contain the temporary data written to and read from memory bythe microprocessor.

[0327] Referring to FIG. 9, the present invention includes a RFTransmitter/Receiver 950 and power amplifier section 960. Referring toFIG. 22, the RF Transmitter/Receiver 950 is divided into three sections:the transmitter module 2201, the receiver module 2202, and the FrequencySynthesizer 2203. Frequency Synthesizer 2203 produces a transmit carrierfrequency TFREQ and a receive carrier frequency RFREQ in response to aFrequency control signal FREQCTRL received from the WAC 920 on theserial link 912. In the transmitter module 2201, the input analog I andQ data signals TXI and TXQ from the VDC are applied to the Quadraturemodulator 2220, which also receives a transmit carrier frequency signalTFREQ from the Frequency Synthesizer 2203 to produce a quadraturemodulated transmit carrier signal TX. The analog transmit carriermodulated signal, an upconverted RF signal, TX is then applied to theTransmit Power Amplifier 2252 of the Power Amplifier 960. The amplifiedtransmit carrier signal is then passed through the High Power PassiveComponents (HPPC) 2253 to the Antenna 2250, which transmits theupconverted RF signal to the communication channel as a CDMA RF signal.In one embodiment of the invention, the Transmit Power Amplifier 2252comprises eight amplifiers of approximately 60 watts peak-to-peak each.

[0328] The HPPC 2253 comprises a lightning protector, an output filter,a 10 dB directional coupler, an isolator, and a high power terminationattached to the isolator.

[0329] A receive CDMA RF signal is received at the antenna 2250 from theRF channel and passed through the HPPC 2253 to the Receive PowerAmplifier 2251. The receive power amplifier 2251 includes, for example,a 30 watt power transistor driven by a 5 watt transistor. The RF receivemodule 2202 has quadrature modulated receive carrier signal RX from thereceive power amplifier. The receive module 2202 includes a Quadraturedemodulator 2210 which takes the receive carrier modulated signal RX andthe receive carrier frequency signal RFREQ from the FrequencySynthesizer 2203, synchronously demodulates the carrier, and providesanalog I and Q channels. These channels are filtered to produce thesignals RXI and RXQ, which are transferred to the VDC 940.

[0330] The Subscriber Unit

[0331]FIG. 23 shows the Subscriber Unit (SU) of one embodiment of thepresent invention. As shown, the SU includes an RF section 2301including a RF modulator 2302, RF demodulator 2303, andsplitter/isolator 2304 which receive Global and Assigned logicalchannels including traffic and control messages and Global Pilot signalsin the Forward link CDMA RF channel signal, and transmit AssignedChannels and Reverse Pilot signals in the Reverse Link CDMA RF channel.The Forward and Reverse links are received and transmitted respectivelythrough antenna 2305. The RF section employs, in one exemplaryembodiment, a conventional dual conversion superheterodyne receiverhaving a synchronous demodulator responsive to the signal ROSC.Selectivity of such a receiver is provided by a 70 MHz transversal SAWfilter (not shown). The RF modulator includes a synchronous modulator(not shown) responsive to the carrier signal TOSC to produce aquadrature modulated carrier signal. This signal is stepped up infrequency by an offset mixing circuit (not shown).

[0332] The SU further includes a Subscriber Line Interface 2310,including the functionality of a control (CC) generator, a DataInterface 2320, an ADPCM encoder 2321, an ADPCM decoder 2322, an SUcontroller 2330, an SU clock signal generator 2331, memory 2332, and aCDMA modem 2340, which is essentially the same as the CDMA modem 1210described above with reference to FIG. 13. It is noted that datainterface 2320, ADPCM Encoder 2321 and ADPCM Decoder 2322 are typicallyprovided as a standard ADPCM Encoder/Decoder chip.

[0333] The Forward Link CDMA RF Channel signal is applied to the RFdemodulator 2303 to produce the Forward link CDMA signal. The ForwardLink CDMA signal is provided to the CDMA modem 2340, which acquiressynchronization with the Global pilot signal, produces global pilotsynchronization signal to the Clock 2331, to generate the system timingsignals, and despreads the plurality of logical channels. The CDMA modem2340 also acquires the traffic messages RMESS and control messages RCTRLand provides the traffic message signals RMESS to the Data Interface2320 and receive control message signals RCTRL to the SU Controller2330.

[0334] The receive control message signals RCTRL include a subscriberidentification signal, a coding signal, and bearer modification signals.The RCTRL may also include control and other telecommunication signalinginformation. The receive control message signal RCTRL is applied to theSU controller 2330, which verifies that the call is for the SU from theSubscriber identification value derived from RCTRL. The SU controller2330 determines the type of user information contained in the trafficmessage signal from the coding signal and bearer rate modificationsignal. If the coding signal indicates the traffic message is ADPCMcoded, the traffic message RVMESS is sent to the ADPCM decoder 2322 bysending a select message to the Data Interface 2320. The SU controller2330 outputs an ADPCM coding signal and bearer rate signal derived fromthe coding signal to the ADPCM decoder 2322. The traffic message signalRVMESS is the input signal to the ADPCM decoder 2322, where the trafficmessage signal is converted to a digital information signal RINF inresponse to the values of the input ADPCM coding signal.

[0335] If the SU controller 2330 determines the type of user informationcontained in the traffic message signal from the coding signal is notADPCM coded, then RDMESS passes through the ADPCM encoder transparently.The traffic message RDMESS is transferred from the Data Interface 2320directly to the Interface Controller (IC) 2312 of the subscriber lineinterface 2310.

[0336] The digital information signal RINF or RDMESS is applied to thesubscriber line interface 2310, including a interface controller (IC)2312 and Line Interface (LI) 2313. For the exemplary embodiment the ICis an Extended PCM Interface Controller (EPIC) and the LI is aSubscriber Line Interface Circuit (SLIC) for POTS which corresponds toRINF type signals, and a ISDN Interface for ISDN which corresponds toRDMESS type signals. The EPIC and SLIC circuits are well known in theart. The subscriber line interface 2310 converts the digital informationsignal RINF or RDMESS to the user defined format. The user definedformat is provided to the IC 2312 from the SU Controller 2330. The LI2310 includes circuits for performing such functions as A-law or μ-lawconversion, generating dial tone and, and generating or interpretingsignaling bits. The line interface also produces the user informationsignal to the SU User 2350 as defined by the subscriber line interface,for example POTS voice, voiceband data or ISDN data service.

[0337] For a Reverse Link CDMA RF Channel, a user information signal isapplied to the LI 2313 of the subscriber line interface 2310, whichoutputs a service type signal and an information type signal to the SUcontroller. The IC 2312 of the subscriber line interface 2310 produces adigital information signal TINF which is the input signal to the ADPCMencoder 2321 if the user information signal is to be ADPCM encoded, suchas for POTS service. For data or other non-ADPCM encoded userinformation, the IC 2312 passes the data message TDMESS directly to theData Interface 2320. The Call control module (CC), including in thesubscriber line interface 2310, derives call control information fromthe User information signal, and passes the call control informationCCINF to the SU controller 2330. The ADPCM encoder 2321 also receivescoding signal and bearer modification signals from the SU controller2330 and converts the input digital information signal into the outputmessage traffic signal TVMESS in response to the coding and bearermodification signals. The SU controller 2330 also outputs the reversecontrol signal which includes the coding signal call controlinformation, and bearer channel modification signal, to the CDMA modem.The output message signal TVMESS is applied to the Data Interface 2320.The Data Interface 2320 sends the user information to the CDMA modem2340 as transmit message signal TMESS. The CDMA modem 2340 spreads theoutput message and reverse control channels TCTRL received from the SUcontroller 2330, and produces the reverse link CDMA Signal. The ReverseLink CDMA signal is provided to the RF transmit section 2301 andmodulated by the RF modulator 2302 to produce the output Reverse LinkCDMA RF channel signal transmitted from antenna 2305.

[0338] SU Controller 2330 receives data RFDAT from the RF Demodulator2303 and RF Modulator 2302 concerning operating characteristics of theRF section 2301, including, for example, measurements of signal gain,signal power, frequency shift. In response to the data RFDAT, SUController 2330 may adjust programmable operating parameters in RFsection 2301.

[0339] In another embodiment of the present invention, the memory 2332is composed of two memory components: a first memory for containing aprogram for loading and use by the SU Controller 2330, and a secondmemory for writing and storing information during operation. The firstmemory may be a programmable memory, such as a FLASH memory. SUController 2330 may receive a new program transmitted to the SU from theCDMA modem 2340 or from an external device (not shown). Upon receivingthe new program, the SU Controller may store the new program in thesecond memory, determine that the program has been correctly received,store the program in the first memory by reprogramming the first memory,and then reboot and load the new software.

[0340] Also, an optional interface to an optional monitoring device 2352may be provided by the SU Controller 2330. SU Controller 2330 mayreceive data MODAT from the CDMA Modem 2340 which may indicate currentvalues of system parameters, such as, for example, system noiseinterference levels, number of established calls, forward and reversepower control parameters, time to access a channel, time to establish achannel, and number of dropped calls. SU Controller 2330 may collect andstore this information in memory 2332 and provide the information to theoptional monitor 2352 if prompted by a user or automatically.

[0341] Call Connection and Establishment Procedure

[0342] The process of bearer channel establishment consists of twoprocedures: the call connection process for a call connection incomingfrom a remote call processing unit such as an RDU (Incoming CallConnection), and the call connection process for a call outgoing fromthe SU (Outgoing Call Connection). Before any bearer channel can beestablished between an RCS and a SU, the SU must register its presencein the network with the remote call processor such as the RDU. When theoff-hook signal is detected by the SU, the SU not only begins toestablish a bearer channel; but also initiates the procedure for an RCSto obtain a terrestrial link between the RCS and the remote processor.As incorporated herein by reference, the process of establishing the RCSand RDU connection is detailed in the DECT V5.1 standard.

[0343] For the Incoming Call Connection procedure shown in FIG. 24,first 2401, the WAC 920 (shown in FIG. 9) receives, via one of the MUXs905, 906 and 907, an incoming call request from a remote call processingunit. This request identifies the target SU and that a call connectionto the SU is desired. The WAC periodically outputs the SBCH channel withpaging indicators for each SU and periodically outputs the FBCH trafficlights for each access channel. In response to the incoming callrequest, the WAC, at step 2420, first checks to see if the identified SUis already active with another call. If so, the WAC returns a busysignal for the SU to the remote processing unit through the MUX,otherwise the paging indicator for the channel is set.

[0344] Next, at step 2402, the WAC checks the status of the RCS modemsand, at step 2421, determines whether there is an available modem forthe call. If a modem is available, the traffic lights on the FBCHindicate that one or more AXCH channels are available. If no channel isavailable after a certain period of time, then the WAC returns a busysignal for the SU to the remote processing unit through the MUX. If anRCS modem is available and the SU is not active (in Sleep mode), the WACsets the paging indicator for the identified SU on the SBCH to indicatean incoming call request. Meanwhile, the access channel modemscontinuously search for the Short Access Pilot signal (SAXPT) of the SU.

[0345] At step 2403, an SU in Sleep mode periodically enters awake mode.In awake mode, the SU modem synchronizes to the Downlink Pilot signal,waits for the SU modem AMF filters and phase locked loop to settle, andreads the paging indicator in the slot assigned to it on the SBCH todetermine if there is a call for the SU 2422. If no paging indicator isset, the SU halts the SU modem and returns to sleep mode. If a pagingindicator is set for an incoming call connection, the SU modem checksthe service type and traffic lights on FBCH for an available AXCH.

[0346] Next, at step 2404, the SU modem selects an available AXCH andstarts a fast transmit power ramp-up on the corresponding SAXPT. For aperiod the SU modem continues fast power ramp-up on SAXPT and the accessmodems continue to search for the SAXPT.

[0347] At step 2405, the RCS modem acquires the SAXPT of the SU andbegins to search for the SU LAXPT. When the SAXPT is acquired, the modeminforms the WAC controller, and the WAC controller sets the trafficlights corresponding to the modem to “red” to indicate the modem is nowbusy. The traffic lights are periodically output while continuing toattempt acquisition of the LAXPT.

[0348] The SU modem monitors, at step 2406, the FBCH AXCH traffic light.When the AXCH traffic light is set to red, the SU assumes the RCS modemhas acquired the SAXPT and begins transmitting LAXPT. The SU modemcontinues to ramp-up power of the LAXPT at a slower rate until Sync-Indmessages are received on the corresponding CTCH. If the SU is mistakenbecause the traffic light was actually set in response to another SUacquiring the AXCH, the SU modem times out because no Sync-Ind messagesare received. The SU randomly waits a period of time, picks a new AXCHchannel, and steps 2404 and 2405 are repeated until the SU modemreceives Sync-Ind messages. Details of the power ramp up method used inthe exemplary embodiment of this invention may be found in the U.S.patent application entitled METHOD OF CONTR0LLING INITIAL POWER RAMP-UPIN CDMA SYSTEMS BY USING SHORT CODES filed on even date herewith, whichis hereby incorporated by reference.

[0349] Next, at step 2407, the RCS modem acquires the LAXPT of the SUand begins sending Sync-Ind messages on the corresponding CTCH. Themodem waits 10 msec for the Pilot and AUX Vector correlator filters andPhase locked loop to settle, but continues to send Synch-Ind messages onthe CTCH. The modem then begins looking for a request message for accessto a bearer channel (MAC_ACC_REQ), from the SU modem.

[0350] The SU modem, at step 2408, receives the Sync-Ind message andfreezes the LAXPT transmit power level. The SU modem then begins sendingrepeated request messages for access to a bearer traffic channel(MAC_ACC_REQ) at fixed power levels, and listens for a requestconfirmation message (MAC_BEARER_CFM) from the RCS modem.

[0351] Next, at step 2409, the RCS modem receives a MAC_ACC_REQ message;the modem then starts measuring the AXCH power level, and starts the APCchannel. The RCS modem then sends the MAC_BEARER_CFM message to the SUand begins listening for the acknowledgment MAC_BEARER_CFM_ACK of theMAC_BEARER_CFM message.

[0352] At step 2410, the SU modem receives the MAC_BEARER_CFM messageand begins obeying the APC power control messages. The SU stops sendingthe MAC_ACC_REQ message and sends the RCS modem the MAC_BEARER_CFM_ACKmessage. The SU begins sending the null data on the AXCH. The SU waits10 msec for the uplink transmit power level to settle.

[0353] The RCS modem, at step 2411, receives the MAC_BEARER_CFM_ACKmessage and stops sending the MAC_BEARER_CFM messages. APC powermeasurements continue.

[0354] Next, at step 2412, both the SU and the RCS modems havesynchronized the sub-epochs, obey APC messages, measure receive powerlevels, and compute and send APC messages. The SU waits 10 msec fordownlink power level to settle.

[0355] Finally, at step 2413, Bearer channel is established andinitialized between the SU and RCS modems. The WAC receives the bearerestablishment signal from the RCS modem, re-allocates the AXCH channeland sets the corresponding traffic light to green.

[0356] For the Outgoing Call Connection shown in FIG. 25, the SU isplaced in active mode by the off-hook signal at the user interface atstep 2501.

[0357] Next, at step 2502, the RCS indicates available AXCH channels bysetting the respective traffic lights.

[0358] At step 2503, the SU synchronizes to the Downlink Pilot, waitsfor the SU modem Vector correlator filters and phase lock loop tosettle, and the SU checks service type and traffic lights for anavailable AXCH.

[0359] Steps 2504 through 2513 are identical to the procedure steps 2404through 2413 for the Incoming Call Connection procedure of FIG. 24, andso are not explained in detail.

[0360] In the previous procedures for Incoming Call Connection andOutgoing Call Connection, the power Ramping-Up process consists of thefollowing events. The SU starts from very low transmit power andincreases its power level while transmitting the short code SAXPT; oncethe RCS modem detects the short code it turns off the traffic light.Upon detecting the changed traffic light, the SU continues ramping-up ata slower rate this time sending the LAXPT. Once the RCS modem acquiresthe LAXPT and sends a message on CTCH to indicate this, the SU keeps itstransmit (TX) power constant and sends the MAC-Access-Request message.This message is answered with a MAC_BEARER_CFM message on the CTCH. Oncethe SU receives the MAC_BEAER_CFM message it switches to the trafficchannel (TRCH) which is the dial tone for POTS.

[0361] When the SU captures a specific user channel AXCH, the RCSassigns a code seed for the SU through the CTCH. The code seed is usedby the spreading code generator in the SU modem to produce the assignedcode for the reverse pilot of the subscriber, and the spreading codesfor associated channels for traffic, call control, and signaling. The SUreverse pilot spreading code sequence is synchronized in phase to theRCS system Global Pilot spreading code sequence, and the traffic, callcontrol, and signaling spreading codes are synchronized in phase to theSU reverse pilot spreading code sequence.

[0362] If the Subscriber unit is successful in capturing a specific userchannel, the RCS establishes a terrestrial link with the remoteprocessing unit to correspond to the specific user channel. For the DECTV5. 1 standard, once the complete link from the RDU to the LE isestablished using the V5.1 ESTABLISHMENT message, a corresponding V5.1ESTABLISHMENT ACK message is returned from the LE to the RDU, and theSubscriber Unit is sent a CONNECT message indicating that thetransmission link is complete.

[0363] Support of Special Service Types

[0364] The system of the present invention includes a bearer channelmodification feature which allows the transmission rate of the userinformation to be switched from a lower rate to a maximum of 64 kb/s.The Bearer Channel Modification (BCM) method is used to change a 32 kb/sADPCM channel to a 64 kb/s PCM channel to support high speed data andfax communications through the spread-spectrum communication system ofthe present invention. Additional details of this technique aredescribed in U.S. Patent Application entitled “CDMA COMMUNICATION SYSTEMWHICH SELECTIVELY SUPPRESSES DATA TRANSMISSIONING DURING ESTABLISHMENTOF A COMMUNICATION CHANNEL” filed on even date herewith and incorporatedherein by reference.

[0365] The system of the present invention includes at least twodifferent types of bearer modification: a first type which uses twolinks set-up over the RF interface and includes an asynchronousswitchover in the transmit and receive directions, and a second typewhich uses only a single link over the RF interface and may includeeither a synchronous or an asynchronous switchover in the transmit andreceive directions. In the following, the term “standard bearer channel”is used to refer to a bearer channel having a transmission rate normallyemployed (for example, 32 kb/sec or encoded channel) and the term “highrate bearer channel” is used to refer to a bearer channel which cansupport a higher transmission rate (for example, 64 kb/sec or decodedchannel). Also, the following uses the term “terrestrial link” toindicate a transmission link between the RCS and a remote processingunit. While the term terrestrial link may commonly refer to a standardtwisted pair transmission link, such as a 1.544 Mb/s T1 or 2.048 Mb/sE1, the terrestrial link contemplated for use with the exemplaryembodiment of the invention is not so limited and can be of any type oftransmission link, having any rate, wired or wireless.

[0366] In the first type of bearer modification, the following describesthe uplink (SU to RCS) process. The standard bearer channel on the RFinterface is established between the RCS and SU, and a correspondinglink exists between the RCS terrestrial interface and the remoteprocessing unit, such as an RDU. The digital transmission rate of thelink between the RCS and remote processing unit normally corresponds toa data encoded rate, which may be, for example, ADPCM at 32 kb/s. TheWAC controller of the RCS monitors the encoded digital data informationof the link received by the Line Interface of the MUX. If the WACcontroller detects the presence of a 2100 Hz tone in the digital data,the WAC first determines whether a high rate channel is available overthe RF interface and, if so, instructs the SU through the assignedlogical control channel to begin bearer modification. The instructioncauses a high rate link to be established between the RCS modem and theSU. In addition, the WAC controller instructs the remote processing unitto establish a second link supporting the high rate link between theremote processing unit and the RCS.

[0367] Once the remote processing unit sets up the high rate link to theRCS, the RCS sends a MODIFY TRANSMIT message to the SU, which respondsby transmitting the digital information on the high rate bearer channel.Consequently, for a brief period, the SU transmits digital data overboth the 32 kb/s and the 64 kb/s links to the RCS at the same time.

[0368] Once the high rate bearer link is established, the SU sends aMODIFY TRANSMIT CONFIRM message to the RCS, which then sends a messageto the remote processing unit to indicate the modification is completeand to switch to the high rate terrestrial link. The remote processingunit sends a message to the RCS to confirm that it switched to the highrate link, which causes the WAC controller to switch its transmission tothe high rate link by removing the standard rate terrestrial link, andalso causes the WAC controller to instruct the RCS modem and the SU toterminate and tear down the standard rate bearer channel (e.g. 32 kb/slink).

[0369] The downlink process for the first type of bearer modification issimilar to the uplink process. The standard bearer channel on the RFinterface is established between the RCS and SU, and a correspondinglink exists between the RCS terrestrial interface and the remoteprocessing unit, such as an RDU. The digital transmission rate of thelink between the RCS and remote processing unit normally corresponds toa data encoded rate, which may be, for example, ADPCM at 32 kb/s. TheWAC controller of the RCS monitors the encoded digital data informationof the link received by the Line Interface of the MUX. If the WACcontroller detects the presence of the 2100 Hz tone in the digital data,the WAC first determines whether a high rate channel is available overthe RF interface and, if so, instructs the SU through the assignedlogical control channel to begin bearer modification. The instructioncauses a high rate bearer channel to be established between the RCSmodem and the SU. In addition, the WAC controller instructs the remoteprocessing unit to establish a second link supporting the high rateterrestrial link between the remote processing unit and the RCS.

[0370] Once the remote processing unit sets up the high rate link to theRCS, the remote processing unit sends an indication to the RCS.Consequently, for a brief period, the remote processing unit and the RCSexchange the same data over both the standard (e.g. 32 kb/s) and thehigh rate (e.g. 64 kb/s) terrestrial links. Then, the RCS sends a MODIFYRECEIVE message to the SU which responds to this message by beginningreception of the digital information on the high rate bearer channel.

[0371] Once the high rate bearer link is established, the SU sends aMODIFY RECEIVE CONFIRM message to the RCS, which then sends a message tothe remote processing unit to indicate the modification is complete, andto tear down the standard rate terrestrial link. The remote processingunit sends a message to the RCS to confirm that it switched transmissionto the high rate link, which causes the WAC controller to instruct theRCS modem and the SU to terminate and tear down the standard bearerchannel.

[0372] Another embodiment of the BCM method incorporates a negotiationbetween the external remote processing unit, such as the RDU, and theRCS to allow for redundant channels on the terrestrial interface, whileonly using one bearer channel on the RF interface. The method describedis a synchronous switchover from the standard (e.g. 32 kb/s) link to thehigh rate (e.g. 64 kb/s) link over the air link which takes advantage ofthe fact that the spreading code sequence timing is synchronized betweenthe RCS modem and SU. In addition, the method uses the advantage of atranscoder within the transmit or receive direction of the RCS modem toconvert the digital data from the standard to the high rate whilepassing through the RCS.

[0373] The uplink process is described first. When the WAC controllerdetects the presence of the 2100 Hz tone in the digital data receivedfrom the SU, and determines that a high rate bearer channel over the RFinterface is available, a transcoder in the RCS modem receives andtranscodes the standard rate digital data signal from the SU into a highrate digital data signal for transmission to the remote processing unit.Then, the WAC controller instructs the remote processing unit toestablish a high rate terrstrial link duplex link between the remoteprocessing unit and the RCS and begin receiving the decoded high ratedigital data signal. The remote processing unit does not receive data onboth terrestrial links concurrently. Instead, once the remote processingunit begins receiving the digital data signal on the high rate link, theRCS is informed of the switch. The, the standard bearer channel on theRF interface is modified to be a high rate bearer channel by a MODIFYTRANSMIT and concurrently the transcoder is removed from thetransmission path.

[0374] Thus, the RCS also informs the SU that the standard bearerchannel is being torn down and to switch processing to transmitunencoded 64 kb/s data on the channel. The SU and RCS exchange controlmessages over the bearer control channel of the assigned channel groupto identify and determine the particular subepoch of the bearer channelspreading code sequence within which the SU will begin transmittingunencoded high rate data to the RCS. Once the subepoch is identified,the switch occurs synchronously at the identified subepoch boundary.This synchronous switchover method is more economical of bandwidth inthe RF interface since the system does not need to maintain theadditional bearer capacity for a 64 kb/s link in order to support aswitchover.

[0375] Next, the downlink process is described. The WAC controllerdetects the presence of the 2100 Hz tone in the digital data receivedfrom the remote processing unit, and determines that a high rate bearerchannel over the RF interface is available, a transcoder in the RCSmodem is assigned to a high rate link from the remote processing unit.The transcoder transcodes the high rate digital data signal to bereceived from the remote processing unit into a standard rate digitaldata signal for transmission to SU. Then, the WAC controller instructsthe remote processing unit to establish a high rate terrestrial linkduplex link between the remote processing unit and the RCS and begintransmitting the high rate data on the high rate link. The RCS receivesdigital data on both terrestrial links concurrently. Once the remoteprocessing unit indicates to the RCS it is transmitting digital data onthe high rate link, the RCS modifies the standard bearer channel on theRF interface to be a high rate bearer channel by a MODIFY RECEIVE andconcurrently the transcoder is removed from the transmission path.

[0376] Thus, the RCS also informs the SU that the standard bearerchannel is being torn down and to switch processing to transmitunencoded high rate digital data on the channel. The SU and RCS exchangecontrol messages over the bearer control channel of the assigned channelgroup to identify and determine the particular subepoch of the bearerchannel spreading code sequence within which the RCS will begintransmitting unencoded high rate data to the SU. Once the subepoch isidentified, the switch occurs synchronously at the identified subepochboundary.

[0377] A further exemplary embodiment of the present invention includingthis second type of bearer modification may switch both the uplink anddownlink directions together and synchronously. In these embodiment, thepreviously described uplink and downlink transmission path processes aresuperimposed. The switchover process may use, however, two differentversions of message sequences to accomplish the synchronous switching.

[0378] In the first version, after the RCS has detected the 2100 Hztone, the RCS transmits on both a standard rate terrestrial link and ahigh rate terestrial link to the remote processing unit, and assigns atranscoder to decode the digital data signal received from the SU fortransmission on the high rate terrestrial link in the uplink path. TheRCS also prepares to receive a high rate terrestrial link from theremote processing unit by assigning a second transcoder to encodedigital data from the remote processing unit and by creating a high rateterrestrial link in the downlink path. Then, the RCS sends a message tothe remote processing unit to begin transmission in the downlink path inboth the standard and high rate terrestrial links, and to switchreception from the standard rate terrestrial link to the high rateterrestrial link in the uplink path.

[0379] The remote processing unit sends a message to the RCS to confirmthat the modification has been accomplished, and the RCS then activatesthe transcoder in the downlink path to encode the digital data signalfor transmission on the RF interface to the SU on a standard rate bearerchannel. Once this is accomplished, the RCS and SU negotiate a time toswitch uplink and downlink paths synchronously. At the decided time, thestandard rate bearer channels in the uplink and downlink paths aremodified to be high rate bearer channels, and the standard rateterrestrial links are torn down.

[0380] In the second version, the SU and RCS negotiate a time tosynchronously modify the standard rate bearer to a high rate bearer, andperform the modification in both the uplink and downlink paths,concurrently with the RCS message to remote processing unit to startreceiving in the uplink in the high rate terrestrial link and to starttransmitting on both the standard rate and high rate terrestrial linksin the downlink. Once the remote processing unit gives confirmation ofthis message, the RCS and remote processing unit tear down the standardrate terrestrial links in both uplink and downlink paths.

[0381] As another special service type, the system of the presentinvention includes a method for conserving capacity over the RFinterface for ISDN types of traffic. This conservation occurs while aknown idle bit pattern is transmitted in the ISDN D-channel when no datainformation is being transmitted. The CDMA system of the presentinvention includes a method to prevent transmission of redundantinformation carried on the D-channel of ISDN networks for signalstransmitted through a wireless communication link. The advantage of suchmethod is that it reduces the amount of information transmitted andconsequently the transmit power and channel capacity used by thatinformation. The method is described as it is used in the RCS. In thefirst step, the controller, such as the WAC of the RCS or the SUcontroller of the SU, monitors the output D-channel from the subscriberline interface for a pre-determined channel idle pattern. A delay isincluded between the output of the line interface and the CDMA modem.Once the idle pattern is detected, the controller inhibits thetransmission of the spread message channel through a message included inthe control signal to the CDMA modem. The controller continues tomonitor the output D-channel of the line interface until the presence ofdata information is detected. When data information is detected, thespread message channel is activated. Because the message channel issynchronized to the associated pilot which is not inhibited, thecorresponding CDMA modem of the other end of the communication link doesnot have to reacquire synchronization to the message channel.

[0382] Drop Out Recovery

[0383] The RCS and SU each monitor the CDMA bearer channel signal toevaluate the quality of the CDMA bearer channel connection. Link qualityis evaluated using the sequential probability ratio test (SPRT)employing adaptive quantile estimation. The SPRT process usesmeasurements of the received signal power; and if the SPRT processdetects that the local spreading code generator has lost synchronizationwith the received signal spreading code or if it detects the absence orlow level of a received signal, the SPRT declares loss of lock (LOL).

[0384] When the LOL condition is declared, the receiver modem of eachRCS and SU begins a Z-search of the input signal with the localspreading code generator. Z-search is well known in the art of CDMAspreading code acquisition and detection and is described in DigitalCommunications and Spread Spectrum Systems, by Robert E. Ziemer andRoger L. Peterson, at pages 492-94 which is incorporated herein byreference. The Z-search algorithm of the present invention tests groupsof eight spreading code phases ahead and behind the last known phase inlarger and larger spreading code phase increments.

[0385] During the LOL condition detected by the RCS, the RCS continuesto transmit to the SU on the Assigned Channels, and continues totransmit power control signals to the SU to maintain SU transmit powerlevel. The method of transmitting power control signals is describedbelow. Successful reacquisition desirably takes place within a specifiedperiod of time. If reacquisition is successful, the call connectioncontinues, otherwise the RCS tears down the call connection bydeactivating and deallocating the RCS modem assigned by the WAC, andtransmits a call termination signal to a remote call processor, such asthe RDU, as described previously.

[0386] When the LOL condition is detected by the SU, the SU stopstransmission to the RCS on the Assigned Channels which forces the RCSinto a LOL condition, and starts the reacquisition algorithm. Ifreacquisition is successful, the call connection continues, and if notsuccessful, the RCS tears down the call connection by deactivating anddeallocating the SU modem as described previously.

Power Control

[0387] General

[0388] The power control feature of the present invention is used tominimize the amount of transmit power used by an RCS and the SUs of thesystem, and the power control subfeature that updates transmit powerduring bearer channel connection is defined as automatic power control(APC). APC data is transferred from the RCS to an SU on the forward APCchannel and from an SU to the RCS on the reverse APC channel. When thereis no active data link between the two, the maintenance power control(MPC) subfeature updates the SU transmit power.

[0389] Transmit power levels of forward and reverse assigned channelsand reverse global channels are controlled by the APC algorithm tomaintain sufficient signal power to interference noise power ratio (SIR)on those channels, and to stabilize and minimize system output power.The present invention uses a closed loop power control mechanism inwhich a receiver decides that the transmitter should incrementally raiseor lower its transmit power. This decision is conveyed back to therespective transmitter via the power control signal on the APC channel.The receiver makes the decision to increase or decrease thetransmitter's power based on two error signals. One error signal is anindication of the difference between the measured and desired despreadsignal powers, and the other error signal is an indication of theaverage received total power.

[0390] As used in the described embodiment of the invention, the termnear-end power control is used to refer to adjusting the transmitter'soutput power in accordance with the APC signal received on the APCchannel from the other end. This means the reverse power control for theSU and forward power control for the RCS; and the term far-end APC isused to refer to forward power control for the SU and reverse powercontrol for the RCS (adjusting the opposite end's transmit power).

[0391] In order to conserve power, the SU modem terminates transmissionand powers-down while waiting for a call, defined as the sleep phase.Sleep phase is terminated by an awaken signal from the SU controller.The SU modem acquisition circuit automatically enters the reacquisitionphase, and begins the process of acquiring the downlink pilot, asdescribed previously.

[0392] Closed Loop Power Control Algorithms

[0393] The near-end power control consists of two steps: first, theinitial transmit power is set; and second, the transmit power iscontinually adjusted according to information received from the far-endusing APC.

[0394] For the SU, initial transmit power is set to a minimum value andthen ramped up, for example, at a rate of 1 dB/ms until either a ramp-uptimer expires (not shown) or the RCS changes the corresponding trafficlight value on the FBCH to “red” indicating that the RCS has locked tothe SU's short pilot SAXPT. Expiration of the timer causes the SAXPTtransmission to be shut down, unless the traffic light value is set tored first, in which case the SU continues to ramp-up transmit power butat a much lower rate than before the “red” signal was detected.

[0395] For the RCS, initial transmit power is set at a fixed value,corresponding to the minimum value necessary for reliable operation asdetermined experimentally for the service type and the current number ofsystem users. Global channels, such as Global Pilot or, FBCH, are alwaystransmitted at the fixed initial power, whereas traffic channels areswitched to APC.

[0396] The APC bits are transmitted as one bit up or down signals on theAPC channel. In the described embodiment, the 64 kb/s APC data stream isnot encoded or interleaved.

[0397] Far-end power control consists of the near-end transmitting powercontrol information for the far-end to use in adjusting its transmitpower.

[0398] The APC algorithm causes the RCS or the SU to transmit +1 if thefollowing inequality holds, otherwise −1.

α₁ e ₁−α₂ e ₂>0  (45)

[0399] Here, the error signal e₁ is calculated as

e ₁ =P _(d)−(1+SNR _(REQ))P _(N)  (46)

[0400] where P_(d) is the despread signal plus noise power, P_(N) is thedespread noise power, and SNR_(REQ) is the desired despread signal tonoise ratio for the particular service type; and

e ₂ =P _(r) −P _(o)  (47)

[0401] where Pr is a measure of the received power and Po is theautomatic gain control (AGC) circuit set point. The weights α₁ and α₂ inequation (33) are chosen for each service type and APC update rate.

[0402] Maintenance Power Control

[0403] During the sleep phase of the SU, the interference noise power ofthe CDMA RF channel may change. The present invention includes amaintenance power control feature (MPC) which periodically adjusts theSU's initial transmit power with respect to the interference noise powerof the CDMA channel. The MPC is the process whereby the transmit powerlevel of an SU is maintained within close proximity of the minimum levelfor the RCS to detect the SU's signal. The MPC process compensates forlow frequency changes in the required SU transmit power.

[0404] The maintenance control feature uses two global channels: one iscalled the status channel (STCH) on reverse link, and the other iscalled the check-up channel (CUCH) on forward link. The signalstransmitted on these channels carry no data and they are generated thesame way the short codes used in initial power ramp-up are generated.The STCH and CUCH codes are generated from a “reserved” branch of theglobal code generator.

[0405] The MPC process is as follows. At random intervals, the SU sendsa symbol length spreading code periodically for 3 ms on the statuschannel (STCH). If the RCS detects the sequence, it replies by sending asymbol length code sequence within the next 3 ms on the check-up channel(CUCH). When the SU detects the response from the RCS, it reduces itstransmit power by a particular step size. If the SU does not see anyresponse from the RCS within that 3 ms period, it increases its transmitpower by the step size. Using this method, the RCS response istransmitted at a power level that is enough to maintain a 0.99 detectionprobability at all SU's.

[0406] The rate of change of traffic load and the number of active usersis related to the total interference noise power of the CDMA channel.The update rate and step size of the maintenance power update signal forthe present invention is determined by using queuing theory methods wellknown in the art of communication theory, such as outlined in“Fundamentals of Digital Switching” (Plenum-New York) edited by McDonaldand incorporated herein by reference. By modeling the call originationprocess as an exponential random variable with mean 6.0 mins, numericalcomputation shows the maintenance power level of a SU should be updatedonce every 10 seconds or less to be able to follow the changes ininterference level using 0.5 dB step size. Modeling the call originationprocess as a Poisson random variable with exponential interarrivaltimes, arrival rate of 2×10⁻⁴ per second per user, service rate of{fraction (1/360)} per second, and the total subscriber population is600 in the RCS service area also yields by numerical computation that anupdate rate of once every 10 seconds is sufficient when 0.5 dB step sizeis used.

[0407] Maintenance power adjustment is performed periodically by the SUwhich changes from sleep phase to awake phase and performs the MPCprocess. Consequently, the process for the MPC feature is shown in FIG.26 and is as follows: First, at step 2601, signals are exchanged betweenthe SU and the RCS maintaining a transmit power level that is close tothe required level for detection: the SU periodically sends a symbollength spreading code in the STCH, and the RCS periodically sends asymbol length spreading code in the CUCH as response.

[0408] Next, at step 2602, if the SU receives a response within 3 msafter the STCH message it sent, it decreases its transmit power by aparticular step size at step 2603; but if the SU does not receive aresponse within 3 ms after the STCH message, it increases its transmitpower by the same step size at step 2604.

[0409] The SU waits, at step 2605, for a period of time before sendinganother STCH message, this time period is determined by a random processwhich averages 10 seconds.

[0410] Thus, the transmit power of the STCH messages from the SU isadjusted based on the RCS response periodically, and the transmit powerof the CUCH messages from the RCS is fixed.

[0411] In an alternative embodiment of the present invention, a slightlydifferent method of maintenence power control is performed in which thebase station actually calculates the power of the received messagesignal from a SU and transmits a message to the SU for power adjustment.This process is similar to the transmit power initialization processprior to call establishment, as described previously.

[0412] The SU awakes from the sleep phase and initially transmits amessage to the base station. The initial transmit power is set to aminimum value and then ramped up, for example, at a rate of 1 dB/msuntil either a ramp-up timer (not shown) expires or the RCS changes thecorresponding traffic light value on the FBCH to “red” indicating thatthe RCS has locked to the SU's short pilot SAXPT. Expiration of thetimer causes the SAXPT transmission to be shut down, unless the trafficlight value is set to red first, in which case the SU continues toramp-up transmit power but at a much lower rate than before the “red”signal was detected.

[0413] For the RCS, initial transmit power is set at a fixed value,corresponding to the minimum value necessary for reliable operation asdetermined experimentally for the service type and the current number ofsystem users. Global channels, such as Global Pilot or, FBCH, are alwaystransmitted at the fixed initial power, whereas traffic channels areswitched to APC.

[0414] If the RCS detects the message sent by the SU, the RCS measuresthe received power and signal to noise ratio of the received signal anddetermines whether the signal power should be increased (unacceptablebit error rate) or decreased (excessive initial transmit power). The RCSmay then communicate the required adjustment to the SU in one of twomethods.

[0415] In the first method, a measured value is determined, which may bean error value and may include information of total received noise powerat the base station, and this value is communicated to the SU through amessage channel. For this method, the SU then adjusts its transmit powerand returns to the sleep phase.

[0416] In the second method, the RCS again determines a measured value,but instead uses this value to transmit APC data on the APC channel tothe SU. The APC bits are transmitted as one bit up or down signals (+1or −1) on the APC channel to increase or decrease the SU transmit power.The SU then responds to the APC data, which is a string of +1's or astring of −1's, until the RCS measures an acceptable initial transmitpower level. Then the RCS modifies the APC data to be an alternatingstring of +1's and −1's, which indicates the SU must keep the transmitpower near a constant level, which becomes the initial transmit level.At this point, the SU may then return to the sleep phase.

[0417] Mapping of Power Control Signal to Logical Channels for APC

[0418] Power control signals are mapped to specified Logical Channelsfor controlling transmit power levels of forward and reverse assignedchannels. Reverse global channels are also controlled by the APCalgorithm to maintain sufficient signal power to interference noisepower ratio (SIR) on those reverse channels, and to stabilize andminimize system output power. The present invention uses a closed looppower control method in which a receiver periodically decides toincrementally raise or lower the output power of the transmitter at theother end. The method also conveys that decision back to the respectivetransmitter. TABLE 13 APC Signal Channel Assignments Link Channels andCall/Connection Power Control Method Signals Status Initial ValueContinuous Reverse link Being Established as determined by APC bits inAXCH power ramping forward APC AXPT channel Reverse link In-Progresslevel established APC bits in APC, OW, during call set-up forward APCTRCH, channel pilot signal Forward link In-Progress fixed value APC bitsin APC, OW, reverse APC TRCH channel

[0419] Forward and reverse links are independently controlled. For acall/connection in process, forward link (TRCHs APC, and OW) power iscontrolled by the APC bits transmitted on the reverse APC channel.During the call/connection establishment process, reverse link (AXCH)power is also controlled by the APC bits transmitted on the forward APCchannel. Table 13 summarizes the specific power control methods for thecontrolled channels.

[0420] The required SIRs of the assigned channels TRCH, APC and OW andreverse assigned pilot signal for any particular SU are fixed inproportion to each other and these channels are subject to nearlyidentical fading, therefore, they are power controlled together.

[0421] Adaptive Forward Power Control

[0422] The AFPC process attempts to maintain the minimum required SIR onthe forward channels during a call/connection. The AFPC recursiveprocess, shown in FIG. 27, consists of the steps of having an SU formthe two error signals e₁ and e₂ in step 2701 where

e ₁ =P _(d)−(1+SNR _(REQ))P _(N)  (36)

e ₂ =P _(r) −P _(o)  (37)

[0423] and P_(d) is the despread signal plus noise power, P_(N) is thedespread noise power, SNR_(REQ) is the required signal to noise ratiofor the service type, P_(r) is a measure of the total received power,and P_(o) is the AGC set point. Next, the SU modem forms the combinederror signal α₁e₁+α₂e₂ in step 2702. Here, the weights L₁ and α₂ arechosen for each service type and APC update rate. In step 2703, the SUhard limits the combined error signal and forms a single APC bit. The SUtransmits the APC bit to the RCS in step 2704 and RCS modem receives thebit in step 2705. The RCS increases or decreases its transmit power tothe SU in step 2706 and the algorithm repeats starting from step 2701.

[0424] For the exemplary embodiment, the inventors have determined thatthe value for the noise power P_(N) may be sampled and averaged over atleast one data symbol, and for greater accuracy may be sampled andaveraged over several symbols. In addition, the inventor's havedetermined that the calculation of the error in equation 36 may have abias making it desireable to adjust the value P_(N) by a constant value.

[0425] Adaptive Reverse Power Control

[0426] The ARPC process maintains the minimum desired SIR on the reversechannels to minimize the total system reverse output power, during bothcall/connection establishment and while the call/connection is inprogress. The recursive ARPC process, shown in FIG. 28, begins at step2801 where the RCS modem forms the two error signals e₁ and e₂ in step2801 where

e ₁ =P _(d)−(1+SNR _(REQ))P _(N)  (38)

e ₂ =P _(rt) −P _(o)  (39)

[0427] and P_(d) is the despread signal plus noise power, P_(N) is thedespread noise power, SNR_(REQ) is the desired signal to noise ratio forthe service type, P_(rt) is a measure of the average total powerreceived by the RCS, and P_(o) is the AGC set point. The RCS modem formsthe combined error signal α₁e₁+α₂e₂ in step 2802 and hard limits thiserror signal to determine a single APC bit in step 2803. The RCStransmits the APC bit to the SU in step 2804, and the bit is received bythe SU in step 2805. Finally, the SU adjusts its transmit poweraccording to the received APC bit in step 2806, and the algorithmrepeats starting from step 2801.

[0428] For the exemplary embodiment, the inventors have determined thatthe value for the noise power P_(N) may be sampled and averaged over atleast one data symbol, and for greater accuracy may be sampled andaveraged over several symbols. In addition, the inventor's havedetermined that the calculation of the error in equation 38 may have abias making it desireable to adjust the value P_(N) by a constant value.TABLE 14 Symbols/Thresholds Used for APC Computation Call/ ConnectionSymbol (and Threshold) Used for Service or Call Type Status APC DecisionDon't care Being AXCH Established ISDN D SU In-Progress one 1/64-kb/ssymbol from TRCH (ISDN-D) ISDN 1B + D SU In-Progress TRCH (ISDN-B) ISDN2B + D SU In-Progress TRCH (one ISDN-B) POTS SU (64 KBPS In-Progress one1/64-KBPS symbol from TRCH, PCM) use 64 KBPS PCM threshold POTS SU (32KBPS In-Progress one 1/64-KBPS symbol from TRCH, ADPCM) use 32 KBPSADPCM threshold Silent Maintenance In-Progress OW (continuous during aCall (any SU) maintenance call)

[0429] SIR and Multiple Channel Types

[0430] The required SIR for channels on a link is a function of channelformat (e.g. TRCH, OW), service type (e.g. ISDN B, 32 KBPS ADPCM POTS),and the number of symbols over which data bits are distributed (e.g. two64 kb/s symbols are integrated to form a single 32 kb/s ADPCM POTSsymbol). Despreader output power corresponding to the required SIR foreach channel and service type is predetermined. While a call/connectionis in progress, several user CDMA logical channels are concurrentlyactive; each of these channels transfers a symbol every symbol period.The SIR of the symbol from the nominally highest SIR channel ismeasured, compared to a threshold and used to determine the APC stepup/down decision each symbol period. Table 14 indicates the symbol (andthreshold) used for the APC computation by service and call type.

[0431] APC Parameters

[0432] APC information is always conveyed as a single bit ofinformation, and the APC Data Rate is equivalent to the APC Update Rate.The APC update rate is 64 kb/s. This rate is high enough to accommodateexpected Rayleigh and Doppler fades, and allow for a relatively high(˜0.2) Bit Error Rate (BER) in the Uplink and Downlink APC channels,which minimizes capacity devoted to the APC.

[0433] The power step up/down indicated by an APC bit is nominallybetween 0.1 and 0.01 dB. The dynamic range for power control is 70 dB onthe reverse link and 12 dB on the forward link for the exemplaryembodiment of the present system.

[0434] An Alternative Embodiment of Multiplexing of APC Information

[0435] The dedicated APC and OW logical channels described previouslycan also be multiplexed together in one logical channel. The APCinformation is transmitted at 64 kb/s. continuously whereas the OWinformation occurs in data bursts. The alternative multiplexed logicalchannel includes the unencoded, non-interleaved 64 kb/s. APC informationon, for example, the In-phase channel and the OW information on theQuadrature channel of the QPSK signal.

[0436] Closed Loop Power Control Implementation

[0437] The closed loop power control during a call connection respondsto two different variations in overall system power. First, the systemresponds to local behavior such as changes in power level of an SU, andsecond, the system responds to changes in the power level of the entiregroup of active users in the system.

[0438] The Power Control system of the exemplary embodiment of thepresent invention is shown in FIG. 29. As shown, the circuitry used toadjust the transmitted power is similar for the RCS (shown as the RCSpower control module 2901) and SU (shown as the SU power control module2902). Beginning with the RCS power control module 2901, the reverselink RF channel signal is received at the RF antenna and demodulated toproduce the reverse CDMA signal RMCH. The signal RMCH is applied to thevariable gain amplifier (VGA1) 2910 which produces an input signal tothe Automatic Gain Control (AGC) Circuit 2911. The AGC 2911 produces avariable gain amplifier control signal into the VGA1 2910. This signalmaintains the level of the output signal of VGA1 2910 at a near constantvalue. The output signal of VGA1 is despread by thedespread-demultiplexer (demux) 2912, which produces a despread usermessage signal MS and a forward APC bit. The forward APC bit is appliedto the integrator 2913 to produce the Forward APC control signal. TheForward APC control signal controls the Forward Link VGA2 2914 andmaintains the Forward Link RF channel signal at a minimum desired levelfor communication.

[0439] The signal power of the despread user message signal MS of theRCS power module 2901 is measured by the power measurement circuit 2915to produce a signal power indication. The output of the VGA1 is alsodespread by the AUX despreader which despreads the signal by using anuncorrelated spreading code, and hence obtains a despread noise signal.The power measurement of this signal is multiplied by 1 plus the desiredsignal to noise ratio (SNRR) to form the threshold signal S1. Thedifference between the despread signal power and the threshold value S1is produced by the subtracter 2916. This difference is the error signalES1, which is an error signal relating to the particular SU transmitpower level. Similarly, the control signal for the VGA1 2910 is appliedto the rate scaling circuit 2917 to reduce the rate of the controlsignal for VGA1 2910. The output signal of scaling circuit 2917 is ascaled system power level signal SP1. The Threshold Compute logic 2918calculates the System Signal Threshold value SST from the RCS userchannel power data signal RCSUSR. The complement of the Scaled systempower level signal, SP1, and the System Signal Power Threshold value SSTare applied to the adder 2919 which produces second error signal ES2.This error signal is related to the system transmit power level of allactive SUs. The input Error signals ES1 and ES2 are combined in thecombiner 2920 produce a combined error signal input to the deltamodulator (DM1) 2921, and the output signal of the DM1 is the reverseAPC bit stream signal, having bits of value +1 or −1, which for thepresent invention is transmitted as a 64 kb/sec signal.

[0440] The Reverse APC bit is applied to the spreading circuit 2922, andthe output signal of the spreading circuit 2922 is the spread-spectrumforward APC message signal. Forward OW and Traffic signals are alsoprovided to spreading circuits 2923, 2924, producing forward trafficmessage signals 1, 2, . . . N. The power level of the forward APCsignal, the forward OW, and traffic message signals are adjusted by therespective amplifiers 2925, 2926 and 2927 to produce the power leveladjusted forward APC, OW, and TRCH channels signals. These signals arecombined by the adder 2928 and applied to the VAG2 2914, which producesforward link RF channel signal.

[0441] The forward link RF channel signal including the spread forwardAPC signal is received by the RF antenna of the SU, and demodulated toproduce the forward CDMA signal FMCH. This signal is provided to thevariable gain amplifier (VGA3) 2940. The output signal of VGA3 isapplied to the Automatic Gain Control Circuit (AGC) 2941 which producesa variable gain amplifier control signal to VGA3 2940. This signalmaintains the level of the output signal of VGA3 at a near constantlevel. The output signal of VAG3 2940 is despread by the despread demux2942, which produces a despread user message signal SUMS and a reverseAPC bit. The reverse APC bit is applied to the integrator 2943 whichproduces the Reverse APC control signal. This reverse APC control signalis provided to the Reverse APC VGA4 2944 to maintain the Reverse link RFchannel signal at a minimum power level.

[0442] The despread user message signal SUMS is also applied to thepower measurement circuit 2945 producing a power measurement signal,which is added to the complement of threshold value S2 in the adder 2946to produce error signal ES3. The signal ES3 is an error signal relatingto the RCS transmit power level for the particular SU. To obtainthreshold S2, the despread noise power indication from the AUXdespreader is multiplied by 1 plus the desired signal to noise ratioSNRR. The AUX despreader despreads the input data using an uncorrelatedspreading code, hence its output is an indication of the despread noisepower.

[0443] Similarly, the control signal for the VGA3 is applied to the ratescaling circuit to reduce the rate of the control signal for VGA3 inorder to produce a scaled received power level RP1 (see FIG. 29). Thethreshold compute circuit computes the received signal threshold RSTfrom the SU measured power signal SUUSR. The complement of the scaledreceived power level RP1 and the received signal threshold RST areapplied to the adder which produces error signal ES4. This error isrelated to the RCS transmit power to all other SUs. The input errorsignals ES3 and ES4 are combined in the combiner and input to the deltamodulator DM2 2947. The output signal of DM2 2947 is the forward APC bitstream signal, with bits having value of value +1 or −1. In theexemplary embodiment of the present invention, this signal istransmitted as a 64 kb/sec signal.

[0444] The Forward APC bit stream signal is applied to the spreadingcircuit 2948, to produce the output reverse spread-spectrum APC signal.Reverse OW and Traffic signals are also input to spreading circuits2949, 2950, producing reverse OW and traffic message signals 1, 2, . . .N, and the reverse pilot is generated by the reverse pilot generator2951. The power level of the reverse APC message signal, reverse OWmessage signal, reverse pilot, and the reverse traffic message signalsare adjusted by amplifiers 2952, 2953, 2954, 2955 to produce the signalswhich are combined by the adder 2956 and input to the reverse APC VGA42944. It is this VGA4 2944 which produces the reverse link RF channelsignal.

[0445] During the call connection and bearer channel establishmentprocess, the closed loop power control of the present invention ismodified, and is shown in FIG. 30. As shown, the circuits used to adjustthe transmitted power are different for the RCS, shown as the InitialRCS power control module 3001; and for the SU, shown as the Initial SUpower control module 3002. Beginning with the Initial RCS power controlmodule 3001, the reverse link RF channel signal is received at the RFantenna and demodulated producing the reverse CDMA signal IRMCH which isreceived by the first variable gain amplifier (VGA1) 3003. The outputsignal of VGA1 is detected by the Automatic Gain Control Circuit (AGC1)3004 which provides a variable gain amplifier control signal to VGA13003 to maintain the level of the output signal of VAG1 at a nearconstant value. The output signal of VGA1 is despread by the despreaddemultiplexer 3005, which produces a despread user message signal IMS.The Forward APC control signal, ISET, is set to a fixed value, and isapplied to the Forward Link Variable Gain Amplifier (VGA2) 3006 to setthe Forward Link RF channel signal at a predetermined level.

[0446] The signal power of the despread user message signal IMS of theInitial RCS power module 3001 is measured by the power measure circuit3007, and the output power measurement is subtracted from a thresholdvalue S3 in the subtracter 3008 to produce error signal ES5, which is anerror signal relating to the transmit power level of a particular SU.The threshold S3 is calculated by multiplying the despread powermeasurement obtained from the AUX despreader by 1 plus the desiredsignal to noise ratio SNRR. The AUX despreader despreads the signalusing an uncorrelated spreading code, hence its output signal is anindication of despread noise power. Similarly, the VGA1 control signalis applied to the rate scaling circuit 3009 to reduce the rate of theVGA1 control signal in order to produce a scaled system power levelsignal SP2. The threshold computation logic 3010 determines an InitialSystem Signal Threshold value (ISST) computed from the user channelpower data signal (IRCSUSR). The complement of the Scaled system powerlevel signal SP2 and the ISST are provided to the adder 3011 whichproduces a second error signal ES6, which is an error signal relating tothe system transmit power level of all active SUs. The value of ISST isthe desired transmit power for a system having the particularconfiguration. The input Error signals ES5 and ES6 are combined in thecombiner 3012 produce a combined error signal input to the deltamodulator (DM3) 3013. DM3 produces the initial reverse APC bit streamsignal, having bits of value +1 or −1, which in the exemplary embodimentis transmitted as a 64 kb/s signal.

[0447] The Reverse APC bit stream signal is applied to the spreadingcircuit 3014, to produce the initial spread-spectrum forward APC signal.The CTCH information is spread by the spreader 3016 to form the spreadCTCH message signal. The spread APC and CTCH signals are scaled by theamplifiers 3015 and 3017, and combined by the combiner 3018. Thecombined signal is applied to VAG2 3006, which produces the forward linkRF channel signal.

[0448] The forward link RF channel signal including the spread forwardAPC signal is received by the RF antenna of the SU and demodulated toproduce the initial forward CDMA signal (IFMCH) which is applied to thevariable gain amplifier (VGA3) 3020. The output signal of VGA3 isdetected by the Automatic Gain Control Circuit (AGC2) 3021 whichproduces a variable gain amplifier control signal for the VGA3 3020.This signal maintains the output power level of the VGA3 3020 at a nearconstant value. The output signal of VAG3 is despread by the despreaddemultiplexer 3022, which produces an initial reverse APC bit that isdependent on the output level of VGA3. The reverse APC bit is processedby the integrator 3023 to produce the Reverse APC control signal. TheReverse APC control signal is provided to the Reverse APC VGA4 3024 tomaintain Reverse link RF channel signal at a defined power level.

[0449] The global channel AXCH signal is spread by the spreadingcircuits 3025 to provide the spread AXCH channel signal. The reversepilot generator 3026 provides a reverse pilot signal, and the signalpower of AXCH and the reverse pilot signal are adjusted by therespective amplifiers 3027 and 3028. The spread AXCH channel signal andthe reverse pilot signal are summed by the adder 3029 to produce reverselink CDMA signal. The reverse link CDMA signal is received by thereverse APC VGA4 3024, which produces the reverse link RF channel signaloutput to the RF transmitter.

[0450] System Capacity Management

[0451] The system capacity management algorithm of the present inventionoptimizes the maximum user capacity for an RCS area, called a cell. Whenthe SU comes within a certain value of maximum transmit power, the SUsends an alarm message to the RCS. The RCS sets the traffic lights whichcontrol access to the system, to “red” which, as previously described,is a flag that inhibits access by the SU's. This condition remains ineffect until the call to the alarming SU terminates, or until thetransmit power of the alarming SU, measured at the SU, is a value lessthan the maximum transmit power. When multiple SUs send alarm messages,the condition remains in effect until either all calls from alarming SUsterminate, or until the transmit power of the alarming SU, measured atthe SU, is less than the maximum transmit power. An alternativeembodiment monitors the bit error rate measurements from the FECdecoder, and holds the RCS traffic lights at “red” until the bit errorrate is less than a predetermined value.

[0452] The blocking strategy of the present invention includes a methodwhich uses the power control information transmitted from the RCS to anSU, and the received power measurements at the RCS. The RCS measures itstransmit power level, detects that a maximum value is reached, anddetermines when to block new users. An SU preparing to enter the systemblocks itself if the SU reaches the maximum transmit power beforesuccessful completion of a bearer channel assignment.

[0453] Each additional user in the system has the effect of increasingthe noise level for all other users, which decreases the signal to noiseratio (SNR) that each user experiences. The power control algorithmmaintains a desired SNR for each user. Therefore, in the absence of anyother limitations, addition of a new user into the system has only atransient effect and the desired SNR is regained.

[0454] The transmit power measurement at the RCS is done by measuringeither the root mean square (rms) value of the baseband combined signalor by measuring the transmit power of the RF signal and feeding it backto digital control circuits. The transmit power measurement may also bemade by the SUs to determine if the unit has reached its maximumtransmit power. The SU transmit power level is determined by measuringthe control signal of the RF amplifier, and scaling the value based onthe service type, such as POTS, FAX, or ISDN.

[0455] The information that an SU has reached the maximum power istransmitted to the RCS by the SU in a message on the Assigned Channels.The RCS also determines the condition by measuring reverse APC changesbecause, if the RCS sends APC messages to the SU to increase SU transmitpower, and the SU transmit power measured at the RCS is not increased,the SU has reached the maximum transmit power.

[0456] The RCS does not use traffic lights to block new users who havefinished ramping-up using the short codes. These users are blocked bydenying them the dial tone and letting them time out. The RCS sends all1's (go down commands) on the APC Channel to make the SU lower itstransmit power. The RCS also sends either no CTCH message or a messagewith an invalid address which would force the FSU to abandon the accessprocedure and start over. The SU, however, does not start theacquisition process immediately because the traffic lights are red.

[0457] When the RCS reaches its transmit power limit, it enforcesblocking in the same manner as when an SU reaches its transmit powerlimit. The RCS turns off all the traffic lights on the FBCH, startssending all 1 APC bits (go down commands) to those users who havecompleted their short code ramp-up but have not yet been given a dialtone, and either sends no CTCH message to these users or sends messageswith invalid addresses to force them to abandon the access process.

[0458] The self blocking process of the SU is as follows. When the SUstarts transmitting the AXCH, the APC starts its power control operationusing the AXCH and the SU transmit power increases. While the transmitpower is increasing under the control of the APC, it is monitored by theSU controller. If the transmit power limit is reached, the SU abandonsthe access procedure and starts over.

[0459] System Synchronization

[0460] The RCS is synchronized either to the PSTN Network Clock signalthrough one of the Line interfaces, as shown in FIG. 10, or to the RCSsystem clock oscillator, which free-runs to provide a master timingsignal for the system. The Global Pilot Channel, and therefore allLogical channels within the CDMA channel, are synchronized to the systemclock signal of the RCS. The Global Pilot (GLPT) is transmitted by theRCS and defines the timing at the RCS transmitter.

[0461] The SU receiver is synchronized to the GLPT, and so behaves as aslave to the Network Clock oscillator. However, the SU timing isretarded by the propagation delay. In the present embodiment of theinvention, the SU modem extracts a 64 KHz and 8 KHz clock signal fromthe CDMA RF Receive channel, and a PLL oscillator circuit creates 2 MHzand 4 MHz clock signals The SU transmitter and hence the LAXPT or ASPTare slaved to the timing of the SU receiver.

[0462] The RCS receiver is synchronized to the LAXPT or the ASPTtransmitted by the SU, however, its timing may be retarded by thepropagation delay. Hence, the timing of the RCS receiver is that of theRCS transmitter retarded by twice the propagation delay.

[0463] Furthermore, the system can be synchronized via a referencereceived from a Global Positioning System receiver (GPS). In a system ofthis type, a GPS receiver in each RCS provides a reference clock signalto all submodules of the RCS. Because each RCS receives the same timereference from the GPS, all of the system clock signals in all of theRCSs are synchronized.

[0464] The RCS Test Configuration Unit

[0465] The RCS as described previously, and shown in FIG. 9, may beconfigured for test purposes, as is shown in FIG. 31. Test purposes maybe, for example, to initiate a communication link to an SU and measuresystem parameters described previously. Providing such measurements to auser may be useful for optimally setting the various programmableparameters of the RCS, which may provide an improved system performancesince the RCS is generally used in a fixed location.

[0466] The test configuration may not, however, require support of alarge number of users and hence, communication channels. For such aconfiguration, the functionality of the WAC 920 of FIG. 9 may beincorporated into the MIU 931. For such incorporation, the systemcontrol functions of the WAC may be implemented in the MIU controller1230 shown in FIG. 12. If only one MUX 905 is present, the time slotinterchange function of the TSI 1101 is simplified, and may not even berequired. The TSI 1101 may also be incorporated into a speciallyconfigured MUX unit.

[0467] As shown in FIG. 31, the test configuration includes an optionaltelephone line interface 3101; MUX 3102; a master MIU 3131, includingMaster CDMA Modem 934, CDMA modems 3135, 3136, channel distributioncircuit (CDC) 3738 and system controller 3120; optional additional MIUs3132; a VDC 940; RF Transmitter/Receiver 950 and Power Amplifier 960. Inaddition, the test unit includes an Input and Display Device (IDD) 3152which may be, for example, a PC having a customized user interfaceprogram. The operation of the VDC 940, RF Transmitter/Receiver 950 andPower Amplifier 960 is the same as described previously with respect tothe RCS.

[0468] The optional telephone line interface 3101 may be used to providesignalling or other telephony signals, as well as providing an interfacebetween a telephone (analog input) and the transport line having acorresponding format. Consequently, the telephone line interface 3101can receive an analog signal, sample and quantize the signal into adigital signal, and provide the digital signal with associated telephonycontrol signals (e.g. network signalling), as a channel in group ofmultiplexed channels. For convenience, the channel for transmission toan SU is termed a forward channel. Since a telephone call isbidirectional, or an SU may itself establish a communication link withthe telephone, a channel for transmission from the SU to the telephoneis termed a reverse channel. The reverse channel is also provided as oneof a group of multiplexed reverse channels to the telephone lineinterface 3101, which receives telephony control signals from thereverse channel and provides a reverse analog signal to the telephone.

[0469] The MUX 3102 is configured to accept a group of mulitplexeddigital signals having a line format, and to separate the line formatcoding and telephony control signals from the channels. This informationand the channels, each having a digital signal corresponding to a user'scommunication channel, are provided to the Master MIU 3131. Thefunctionality of the MUX is as described previously with reference toFIG. 10.

[0470] The Master MIU 3131, including Master CDMA Modem 934, performsthose functions as described previously with reference to FIG. 12through FIG. 20. However, the MIU 2131 of the Test Configuration maycontain the following modifications: 1) referring to FIG. 12, the PCMinterface 1220 may include additional interface components to providetime slot interchange functionality; and 2) the System Controller 3120of FIG. 31 may be formed from the MIU controller 1230 of FIG. 12 by thefollowing modifications: 1) an optional increase in the capacity ofmemory 1242 for storage of modem data, 2) addition of an I/O interfacefor communication with an external device, 3) addition of a control linkCTRL1 in order to communicate with and supervise optional additional MIU3132 and MIU 3132 modems (not shown), 4) addition of a control linkCTRL1 in order to communicate with and supervise operation of the VDC940, RF Transmitter/Receiver 950 and Power Amplifier 960, and 5) supportof communication channel routing by the CDC 3738.

[0471] The System Controller 3120 may receive data signals MODAT_(k),k=1, 2, . . . N, from the Master CDMA Modem 934 and each of the CDMAModems 3135 and 3136. Each corresponding data signal MODAT_(i) mayindicate current values of system parameters, such as, for example,system noise interference levels, number of established calls, forwardand reverse power control parameters, time to access a channel, time toestablish a channel, and number of dropped calls. System Controller 3120may collect and store this information in memory (not shown) and providethe information to the IDD 3152 through the I/O interface (not shown) ifprompted by a user (or this may occur automatically).

[0472] The IDD 3152 is typically a remote terminal, such as a PC, whichmay have software programs that support a human-machine interface, inwhich commands are parsed and formed into a request to the SystemController 3120 for particular system parameter values. In addition, theIDD 3152 may also have software programs that receive the particularsystem parameter values (as, for example, a digital data signal having apredetermined data record format) and format the data into apredetermined display format. The display format can be, for example,graphs, values, history, or real time display of changing values.

[0473] Although the invention has been described in terms of multipleexemplary embodiments, it is understood by those skilled in the art thatthe invention may be practiced with modifications to the embodimentsthat are within the scope of the invention as defined by the followingclaims.

What is claimed is:
 1. A subscriber unit for use in a multiple accessspread-spectrum communication system, the subscriber unit comprising: aspread spectrum radio interface, responsive to a rate function signalfrom a base station, wherein the base station assigns the rate functionones of a plurality of spread-spectrum message channels; a firstdespreading means for recovering a respective information signal andmodifying signal from a first one of the spread spectrum messagechannels; and information channel mode modification means responsive tothe rate function signal for reassigning an output of the firstdespreading means to a second despreading means which supports thedifferent information signal rate, the second despreading meanscorresponding to a second spread spectrum message channel, whereby thesubscriber unit has a capability of communicating by dynamicallychanging a transmission rate of an information signal transmitted to asubscriber through a first spread-spectrum message channel of theplurality of spread spectrum message channels.
 2. The subscriber unitaccording to claim 1 further comprising: means for synchronizing amessage according to sequential sub-epoch boundaries of the plurality ofspread spectrum message channels; and means for sending the informationsignal through the first spread spectrum message channel prior to one ofthe sub-epoch boundaries and for sending the information signal throughthe second spread spectrum message channel to the substantial exclusionof the first spread spectrum message channel subsequent to the sub-epochboundary.
 3. The subscriber unit according to claim 1 wherein: the ratefunction signal corresponds to a dynamically changed transmission rateof an information signal received from a remote processing unit and anidentification of an information signal rate of the respectiveinformation signal; and the rate function signal comprises a modifysignal corresponding to the information signal rate.
 4. The subscriberunit according to claim 3 comprising the information signal and themodify signal assigned to respectively different ones of the pluralityof spread-spectrum message channels.
 5. The subscriber unit according toclaim 3 comprising: the information signal and the modify signalassigned to respectively different ones of the plurality ofspread-spectrum message channels; and information channel modemodification means responsive to the modify signal for switching therespective information signal from the first spread-spectrum messagechannel supporting a first information channel rate to one otherspread-spectrum message channel, the one other spread-spectrum messagechannel having a different information channel rate supporting theidentified information signal rate.
 6. The subscriber unit according toclaim 3 comprising: the information signal and the modify signalassigned to respectively different ones of the plurality ofspread-spectrum message channels corresponding to a modify signalresponse from the base station; the modify signal corresponding to aninformation signal rate; the rate signal corresponding to an identifiedinformation signal rate of the respective information signal; and theinformation signal and the modify signal assigned to different ones ofthe plurality of spread-spectrum message channels.
 7. A subscriber unitfor use in a multiple access spread-spectrum communication system, thesubscriber unit capable of communicating by dynamically changing atransmission rate of an information signal transmitted to a subscriberthrough a first spread-spectrum message channel of a plurality of spreadspectrum message channels, the subscriber unit comprising: a spreadspectrum radio interface, responsive to a rate function signal from abase station, wherein the base station assigns the rate function ones ofthe plurality of spread-spectrum message channels; a first despreadingcircuit, the first despreading circuit capable of recovering therespective information signal and modifying signal from the first spreadspectrum message channel; and an information channel mode modificationcircuit responsive to the rate function signal for reassigning an outputof the first despreading circuit to a second despreading circuit whichsupports the different information signal rate, the second despreadingcircuit corresponding to a second spread spectrum message channel. 8.The subscriber unit according to claim 7 further comprising: asynchronization circuit, the synchronization circuit capable ofsynchronizing a message according to sequential sub-epoch boundaries ofthe plurality of spread spectrum message channels; and a circuit sendingthe information signal through the first spread spectrum message channelprior to one of the sub-epoch boundaries and for sending the informationsignal through the second spread spectrum message channel to thesubstantial exclusion of the first spread spectrum message channelsubsequent to the sub-epoch boundary.
 9. The subscriber unit accordingto claim 7 wherein: the rate function signal corresponds to adynamically changed transmission rate of an information signal receivedfrom a remote processing unit and an identification of an informationsignal rate of the respective information signal; and the rate functionsignal comprises a modify signal corresponding to the information signalrate.
 10. The subscriber unit according to claim 9 comprising theinformation signal and the modify signal assigned to respectivelydifferent ones of the plurality of spread-spectrum message channels. 11.The subscriber unit according to claim 9 comprising: the informationsignal and the modify signal assigned to respectively different ones ofthe plurality of spread-spectrum message channels; and an informationchannel mode modification circuit responsive to the modify signal forswitching the respective information signal from the firstspread-spectrum message channel supporting a first information channelrate to one other pre-determined spread-spectrum message channel, theone other pre-determined spread-spectrum message channel having adifferent information channel rate supporting the identified informationsignal rate.
 12. The subscriber unit according to claim 9 comprising:the information signal and the modify signal assigned to respectivelydifferent ones of the plurality of spread-spectrum message channelscorresponding to a modify signal response from the base station; themodify signal corresponding to the information signal rate; the ratesignal corresponding to an identified information signal rate of therespective information signal; and the information signal and the modifysignal assigned to different ones of the plurality of spread-spectrummessage channels.
 13. A subscriber unit for use in a multiple accessspread-spectrum communication system, in which a transmission rate of aninformation signal transmitted to a subscriber corresponding to thesubscriber unit has a dynamically changed transmission rate, thesubscriber unit comprising: synchronization circuitry capable ofsynchronizing a message according to sequential sub-epoch boundaries ofthe plurality of spread spectrum message channels; and a spread spectrumradio interface capable of sending the information signal through thefirst spread spectrum message channel prior to one of the sub-epochboundaries and sending the information signal through the second spreadspectrum message channel to the substantial exclusion of the firstspread spectrum message channel subsequent to the sub-epoch boundary.14. The subscriber unit according to claim 13 comprising: the spreadspectrum radio interface responsive to a rate function signal from abase station, the rate function assigned to ones of the plurality ofspread-spectrum message channels; a first despreader, the firstdespreader providing an information signal and modifying signal from thefirst spread spectrum message channel; and information channel signalprocessor, the information channel signal processor modifying theinformation channel signal in response to the rate function signal forreassigning the signal from the first despreader to a second despreader.15. The subscriber unit according to claim 14 wherein the seconddespreader provides an output corresponding to a second spread spectrummessage channel.
 16. The subscriber unit according to claim 13comprising: a circuit receiving a rate function signal which correspondsto a dynamically changed transmission rate of an information signalreceived from a remote processing unit and an identification of aninformation signal rate of the respective information signal, whereinthe rate function signal comprises a modify signal corresponding to theinformation signal rate.
 17. The subscriber unit according to claim 16comprising the information signal and the modify signal assigned torespectively different ones of the plurality of spread-spectrum messagechannels.
 18. The subscriber unit according to claim 16 comprising: theinformation signal and the modify signal assigned to respectivelydifferent ones of the plurality of spread-spectrum message channels; andinformation channel mode modification circuit responsive to the modifysignal for switching the respective information signal from the firstspread-spectrum message channel supporting a first information channelrate to one other pre-determined spread-spectrum message channel, theone other pre-determined spread-spectrum message channel having adifferent information channel rate supporting the identified informationsignal rate.
 19. The subscriber unit according to claim 16 comprising:the information signal and the modify signal assigned to respectivelydifferent ones of the plurality of spread-spectrum message channelscorresponding to a modify signal response from the base station; themodify signal corresponding to the information signal rate; the ratesignal corresponding to an identified information signal rate of therespective information signal; and the information signal and the modifysignal assigned to different ones of the plurality of spread-spectrummessage channels.